LT3475IFE-1#PBF - LINEAR TECHNOLOGY

LT3475/LT3475-1

3475fb

Dual Step-Down

1.5A LED Driver

The LT®3475/LT3475-1 are dual step-down DC/DC

converters designed to operate as a constant-current

source. An internal sense resistor monitors the output

current allowing accurate current regulation ideal for

driving high current LEDs. The high side current sense al-

lows grounded cathode LED operation. High output current

accuracy is maintained over a wide current range, from

50mA to 1.5A, allowing a wide dimming range. Unique

PWM circuitry allows a dimming range of 3000:1, avoid-

ing the color shift normally associated with LED current

dimming.

The high switching frequency offers several advantages,

permitting the use of small inductors and ceramic capaci-

tors. Small inductors combined with the 20 lead TSSOP

surface mount package save space and cost versus

alternative solutions. The constant switching frequency

combined with low-impedance ceramic capacitors result

in low, predictable output ripple.

With its wide input range of 4V to 36V, the LT3475/LT3475-1

regulate a broad array of power sources. A current mode

PWM architecture provides fast transient response and

cycle-by-cycle current limiting. Frequency foldback and

thermal shutdown provide additional protection.

■ Automotive and Avionic Lighting

■ Architectural Detail Lighting

■ Display Backlighting

■ Constant-Current Sources

■ True Color PWMTM Delivers Constant Color with

3000:1 Dimming Range

■ Wide Input Range: 4V to 36V Operating, 40V

Maximum

■ Accurate and Adjustable Control of LED Current

from 50mA to 1.5A

■ High Side Current Sense Allows Grounded Cathode

LED Operation

■ Open LED (LT3475) and Short Circuit Protection

■ LT3475-1 Drives LED Strings Up to 25V

■ Accurate and Adjustable 200kHz to 2MHz

Switching Frequency

■ Anti-Phase Switching Reduces Ripple

■ Uses Small Inductors and Ceramic Capacitors

■ Available in the Compact 20-Lead TSSOP Thermally

Enhanced Surface Mount Package

Dual Step-Down 1.5A LED Driver

APPLICATIO S

FEATURES DESCRIPTIO

TYPICAL APPLICATIO

Efﬁ ciency

, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.

All other trademarks are the property of their respective owners. Patents Pending.

BOOST1 BOOST2

SW1 SW2

OUT1 OUT2

LED1 LED2

PWM1 PWM2

VC1 VC2

REF RT

VADJ1 VADJ2

GND

LT3475

VIN

5V TO 36V

SHDN

0.22μF

4.7μF

10μH 10μH

0.22μF

24.3k

2.2μF 2.2μF

DIMMING*

CONTROL

*DIMMING

CONTROL

0.1μF

3475 TA01

1.5A LED

CURRENT

1.5A LED

CURRENT

fSW = 600kHz*SEE APPLICATIONS SECTION FOR DETAILS

LED CURRENT (A)

EFFICIENCY (%)

3475 TA01b

55 0.5 1

1.5

TWO SERIES CONNECTED

WHITE 1.5A LEDS

SINGLE WHITE 1.5A LED

VIN = 12V

LT3475/LT3475-1

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VIN Pin .........................................................(-0.3V), 40V

BOOST Pin Voltage ...................................................60V

BOOST Above SW Pin ...............................................30V

OUT, LED, Pins (LT3475) ...........................................15V

OUT, LED Pins (LT3475-1) .........................................25V

PWM Pin ...................................................................15V

VADJ Pin ......................................................................6V

VC, RT

, REF Pins ..........................................................3V

SHDN Pin ...................................................................VIN

Maximum Junction Temperature (Note 2)............. 125°C

Operating Temperature Range (Note 3)

LT3475E/LT3475E-1 ............................. –40°C to 85°C

LT3475I/LT3475I-1 ............................. –40°C to 125°C

Storage Temperature Range ................... –65°C to 150°C

Lead Temperature Range (Soldering, 10 sec) ....... 300°C

(Note 1)

The

● denotes the speciﬁ cations which apply over the full operating

temperature range, otherwise speciﬁ cations are at TA = 25°C. VIN = 12V, VBOOST = 16V, VOUT = 4V unless otherwise noted (Note 3)

PARAMETER CONDITIONS MIN TYP MAX UNITS

Minimum Input Voltage ●3.7 4 V

Input Quiescent Current Not Switching 6 8 mA

Shutdown Current

SHDN = 0.3V, VBOOST = VOUT = 0V

0.01 2 μA

ELECTRICAL CHARACTERISTICS

ABSOLUTE AXI U RATI GS

FE PACKAGE

20-LEAD PLASTIC TSSOP

TOP VIEW

OUT1

LED1

BOOST1

SW1

VIN

SW2

BOOST2

LED2

OUT2

PWM1

VADJ1

VC1

REF

SHDN

GND

VC2

VADJ2

PWM2

TJMAX = 125°C, θJA = 30°C/W, θJC = 8°C/W

EXPOSED PAD (PIN 21) IS GROUND AND MUST

BE ELECTRICALLY CONNECTED TO THE PCB.

LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE

LT3475EFE#PBF LT3475EFE#TRPBF LT3475EFE 20-Lead Plastic TSSOP –40°C to 85°C

LT3475IFE#PBF LT3475IFE#TRPBF LT3475IFE 20-Lead Plastic TSSOP –40°C to 125°C

LT3475EFE-1#PBF LT3475EFE-1#TRPBF LT3475FE-1 20-Lead Plastic TSSOP –40°C to 85°C

LT3475IFE-1#PBF LT3475IFE-1#TRPBF LT3475FE-1 20-Lead Plastic TSSOP –40°C to 125°C

Consult LTC Marketing for parts speciﬁ ed with wider operating temperature ranges. *The temperature grade is identiﬁ ed by a label on the shipping container.

Consult LTC Marketing for information on non-standard lead based ﬁ nish parts.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/

For more information on tape and reel speciﬁ cations, go to: http://www.linear.com/tapeandreel/

PIN CONFIGURATION

ORDER INFORMATION

LT3475/LT3475-1

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Note 1: Stresses beyond those listed under Absolute Maximum Ratings

may cause permanent damage to the device. Exposure to any Absolute

Maximum Rating condition for extended periods may affect device

reliability and lifetime.

Note 2: This IC includes overtemperature protection that is intended

to protect the device during momentary overload conditions. Junction

temperature will exceed 125°C when overtemperature protection is active.

Continuous operation above the speciﬁ ed maximum operating junction

temperature may impair device reliability.

Note 3: The LT3475E and LT3475E-1 are guaranteed to meet performance

speciﬁ cations from 0°C to 85°C. Speciﬁ cations over the –40°C to 85°C

operating temperature range are assured by design, characterization and

correlation with statistical process controls. The LT3475I and LT3475I-1

are guaranteed to meet performance speciﬁ cations over the –40°C to

125°C operating temperature range.

Note 4: Current ﬂ ows out of pin.

Note 5: Current ﬂ ows into pin.

Note 6: Current limit is guaranteed by design and/or correlation to static

test. Slope compensation reduces current limit at higher duty cycles.

The ● denotes the speciﬁ cations which apply over the full operating

temperature range, otherwise speciﬁ cations are at TA = 25°C. VIN = 12V, VBOOST = 16V, VOUT = 4V unless otherwise noted (Note 3)

PARAMETER CONDITIONS MIN TYP MAX UNITS

LED Pin Current VADJ Tied to VREF • 2/3

VADJ Tied to VREF • 7/30

LT3475E/LT3475E-1 0°C to 85°C

●

0.97

0.94

0.336

0.325

0.31

1.00

0.350

1.03

1.04

0.364

0.375

0.385

REF Voltage ●1.22 1.25 1.27 V

Reference Voltage Line Regulation 4V < VIN < 40V 0.05 %/V

Reference Voltage Load Regulation 0 < IREF < 500μA 0.0002 %/μA

VADJ Pin Bias Current (Note 4) ●40 400 nA

Switching Frequency RT = 24.3k ●530 600 640 kHz

Maximum Duty Cycle RT = 24.3k

RT = 4.32k

RT = 100k

●90 95

Switching Phase RT = 24.3k 150 180 210 Deg

Foldback Frequency RT = 24.3k, VOUT = 0V 80 kHz

SHDN

Threshold (to Switch) 2.5 2.6 2.74 V

SHDN

Pin Current (Note 5) VSHDN =

2.6V

7911 μA

PWM Threshold 0.3 0.8 1.2 V

VC Switching Threshold 0.8 V

VC Source Current VC = 1V 50 μA

VC Sink Current VC = 1V 50 μA

LED to VC Transresistance 500 V/A

LED to VC Current Gain 1 mA/μA

VC to Switch Current Gain 2.6 A/V

VC Clamp Voltage 1.8 V

VC Pin Current in PWM Mode VC = 1V, VPWM = 0.3V ●10 400 nA

OUT Pin Clamp Voltage (LT3475) 13.5 14 14.5 V

OUT Pin Current in PWM Mode VOUT = 4V, VPWM = 0.3V ●25 50 μA

Switch Current Limit (Note 6) 2.3 2.7 3.2 A

Switch VCESAT ISW =1.5A 350 500 mV

BOOST Pin Current ISW =1.5A 25 40 mA

Switch Leakage Current 0.1 10 μA

Minimum Boost Voltage Above SW 1.8 2.5 V

ELECTRICAL CHARACTERISTICS

LT3475/LT3475-1

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Switch Current Limit

vs Duty Cycle

Switch Current Limit vs

Temperature

Oscillator Frequency

vs Temperature Oscillator Frequency Foldback Oscillator Frequency vs RT

Current Limit vs Output Voltage

TYPICAL PERFOR A CE CHARACTERISTICS

LED Current vs VADJ LED Current vs Temperature Switch On Voltage

VADJ (V)

LED CURRENT (A)

0.25

0.50

0.75

1.00

1.25

1.50

0.25 0.5 0.75 1

3475 G01

1.25

TA = 25°C

TEMPERATURE (˚C)

–50

LED CURRENT (A)

0.8

1.0

1.2

25 75

3475 G02

0.6

0.4

–25 0 50 100 125

0.2

VADJ = VREF • 7/30

VADJ = VREF • 2/3

SWITCH CURRENT (A)

SWITCH ON VOLTAGE (mV)

100

200

300

400

0.5 1.0 1.5 2.0

3475 G03

500

600 TA = 25°C

DUTY CYCLE (%)

CURRENT LIMIT (A)

0.5

1.0

1.5

2.0

2.5

3.0

20 40 60 80

3475 G04

100

MINIMUM

TYPICAL

TA = 25°C

TEMPERATURE (°C)

–50

2.0

2.5

3.5

25 75

3475 G05

1.5

1.0

–25 0 50 100 125

0.5

3.0

CURRENT LIMIT (A)

VOUT (V)

CURRENT LIMIT (A)

0.5

1.0

1.5

2.0

0.5 2.5 3.0 3.52.01.0 1.5 4.0

3475 G06

2.5

3.0 TA = 25°C

TEMPERATURE (˚C)

–50

OSCILLATOR FREQUENCY (kHz)

600

650

700

25 75

3475 G07

550

500

–25 0 50 100 125

450

400

RT = 24.3kΩ

VOUT (V)

500

600

700

2.0

3475 G08

400

300

0.5 1.0 1.5 2.5

200

100

OSCILLATOR FREQUENCY (kHz)

TA = 25°C

RT = 24.3kΩ

RT (kΩ)

OSCILLATOR FREQUENCY (kHz)

1000

10 100

3475 G09

TA = 25°C

LT3475/LT3475-1

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PI FU CTIO S

UUU

Boost Pin Current Quiescent Current

OUT1, OUT2 (Pins 1, 10): The OUT pin is the input to the

current sense resistor. Connect this pin to the inductor

and the output capacitor.

LED1, LED2 (Pins 2, 9): The LED pin is the output of

the current sense resistor. Connect the anode of the LED

here.

VIN (Pins 5, 6): The VIN pins supply current to the internal

circuitry and to the internal power switches and must be

locally bypassed.

SW1, SW2 (Pins 4, 7): The SW pin is the output of the

internal power switch. Connect this pin to the inductor,

switching diode and boost capacitor.

BOOST1, BOOST2 (Pins 3, 8): The BOOST pin is used to

provide a drive voltage, higher than the input voltage, to

the internal bipolar NPN power switch.

GND (Pins 15, Exposed Pad Pin 21): Ground. Tie the GND

pin and the exposed pad directly to the ground plane. The

exposed pad metal of the package provides both electrical

contact to ground and good thermal contact to the printed

circuit board. The exposed pad must be soldered to the

circuit board for proper operation. Use a large ground plane

and thermal vias to optimize thermal performance.

Open-Circuit Output Voltage and

Input Current

Reference Voltage

Minimum Input Voltage, Single

1.5A White LED

Minimum Input Voltage, Two Series

Connected 1.5A White LEDs

TYPICAL PERFOR A CE CHARACTERISTICS

SWITCH CURRENT (A)

BOOST PIN CURRENT (mA)

1.5

3475 G10

0.5 1.0 2.0

35 TA = 25°C

VIN (V)

OUTPUT VOLTAGE (V)

INPUT CURRENT (mA)

3475 G12

010 20 30

TA = 25°C

LT3475-1

LT3475

OUTPUT VOLTAGE

INPUT CURRENT

VIN (V)

INPUT CURRENT (mA)

3475 G11

10 20 40

7TA = 25°C

LED CURRENT (A)

VIN (V)

0.5 1

3475 G14

1.5

TA = 25°C

TO RUN LED VOLTAGE

TO START

TEMPERATURE (˚C)

–50

VREF (V)

1.26

1.27

1.28

25 75

3475 G13

1.25

1.24

–25 0 50 100 125

1.23

1.22

LED CURRENT (A)

VIN (V)

0.5 1

3475 G15

1.5

TA = 25°C

TO RUN

LED VOLTAGE

TO START

LT3475/LT3475-1

3475fb

3475 BD

∑

INT REG

AND

UVLO

SLAVE

OSC

SLOPE COMP SLOPE COMP

MOSC 1 MOSC 2

FREQUENCY

FOLDBACK

FREQUENCY

FOLDBACK

–

∑

MASTER

OSC

SLAVE

OSC

–

DRIVER

gm1 gm2

BOOST1 BOOST2

D1 D2

D3 D4

DLED1 DLED 2

Q1 Q2

Q3 Q4

C1 C2

OUT1

OUT2

ADJ1

ADJ2

SHDN

GND

EXPOSED

PAD

REF

PWM 1

PWM2

LED1 LED2

SW1 SW2

OUT1 OUT2

0.067Ω 100Ω 0.067Ω100Ω

1.25k

1.25V

1.25k

2V 2V

L1 L2

DRIVER

C1 C2

BLOCK DIAGRAM

RT (Pin 14): The RT pin is used to set the internal

oscillator frequency. Tie a 24.3k resistor from RT to GND

for a 600kHz switching frequency.

SHDN

(Pin 16): The

SHDN

pin is used to shut down the

switching regulator and the internal bias circuits. The

2.6V switching threshold can function as an accurate

undervoltage lockout. Pull below 0.3V to shut down the

LT3475/LT3475-1. Pull above 2.6V to enable the LT3475/

LT3475-1. Tie to VIN if the

SHDN

function is unused.

REF (Pin 17): The REF pin is the buffered output of the

internal reference. Either tie the REF pin to the VADJ pin

for a 1.5A output current, or use a resistor divider to

generate a lower voltage at the VADJ pin. Leave this pin

unconnected if unused.

VC1, VC2 (Pins 18, 13): The VC pin is the output of the

internal error amp. The voltage on this pin controls the

peak switch current. Use this pin to compensate the

control loop.

VADJ1, VADJ2 (Pins 19, 12): The VADJ pin is the input to

the internal voltage-to-current ampliﬁ er. Connect the VADJ

pin to the REF pin for a 1.5A output current. For lower

output currents, program the VADJ pin using the following

formula: ILED = 1.5A • VADJ/1.25V.

PWM1, PWM2 (Pins 20, 11): The PWM pin controls the

connection of the VC pin to the internal circuitry. When

the PWM pin is low, the VC pin is disconnected from the

internal circuitry and draws minimal current. If the PWM

feature is unused, leave this pin unconnected.

PI FU CTIO S

UUU

LT3475/LT3475-1

3475fb

The LT3475 is a dual constant frequency, current mode

regulator with internal power switches capable of gen-

erating constant 1.5A outputs. Operation can be best

understood by referring to the Block Diagram.

If the SHDN pin is tied to ground, the LT3475 is shut

down and draws minimal current from the input source

tied to VIN. If the SHDN pin exceeds 1V, the internal bias

circuits turn on, including the internal regulator, reference

and oscillator. The switching regulators will only begin to

operate when the SHDN pin exceeds 2.6V.

The switcher is a current mode regulator. Instead of directly

modulating the duty cycle of the power switch, the feedback

loop controls the peak current in the switch during each

cycle. Compared to voltage mode control, current mode

control improves loop dynamics and provides cycle-by-

cycle current limit.

A pulse from the oscillator sets the RS ﬂ ip-ﬂ op and turns

on the internal NPN bipolar power switch. Current in the

switch and the external inductor begins to increase. When

this current exceeds a level determined by the voltage at

VC, current comparator C1 resets the ﬂ ip-ﬂ op, turning

off the switch. The current in the inductor ﬂ ows through

the external Schottky diode and begins to decrease. The

cycle begins again at the next pulse from the oscillator.

In this way, the voltage on the VC pin controls the current

through the inductor to the output. The internal error

ampliﬁ er regulates the output current by continually

adjusting the VC pin voltage. The threshold for switching

on the VC pin is 0.8V, and an active clamp of 1.8V limits

the output current.

The voltage on the VADJ pin sets the current through the

LED pin. The NPN, Q3, pulls a current proportional to the

voltage on the VADJ pin through the 100Ω resistor. The gm

ampliﬁ er servos the VC pin to set the current through the

0.067Ω resistor and the LED pin. When the voltage drop

across the 0.067Ω resistor is equal to the voltage drop

across the 100Ω resistor, the servo loop is balanced.

Tying the REF pin to the VADJ pin sets the LED pin current

to 1.5A. Tying a resistor divider to the REF pin allows the

programming of LED pin currents of less than 1.5A. LED

pin current can also be programmed by tying the VADJ pin

directly to a voltage source.

An LED can be dimmed with pulse width modulation

using the PWM pin and an external NFET. If the PWM

pin is unconnected or is pulled high, the part operates

nominally. If the PWM pin is pulled low, the VC pin is dis-

connected from the internal circuitry and draws minimal

current from the compensation capacitor. Circuitry draw-

ing current from the OUT pin is also disabled. This way,

the VC pin and the output capacitor store the state of

the LED pin current until the PWM is pulled high again.

This leads to a highly linear relationship between pulse

width and output light, allowing for a large and accurate

dimming range.

The RT pin allows programming of the switching frequency.

For applications requiring the smallest external components

possible, a fast switching frequency can be used. If low

dropout or very high input voltages are required, a slower

switching frequency can be programmed.

During startup VOUT will be at a low voltage. The NPN,

Q3, can only operate correctly with sufﬁ cient voltage

of ≈1.7V at VOUT

, A comparator senses VOUT and forces

the VC pin high until VOUT rises above 2V, and Q3 is op-

erating correctly.

The switching regulator performs frequency foldback

during overload conditions. An ampliﬁ er senses when

VOUT is less than 2V and begins decreasing the oscillator

frequency down from full frequency to 15% of the nominal

frequency when VOUT = 0V. The OUT pin is less than 2V

during startup, short circuit, and overload conditions.

Frequency foldback helps limit switch current under these

conditions.

The switch driver operates either from VIN or from the

BOOST pin. An external capacitor and Schottky diode

are used to generate a voltage at the BOOST pin that

is higher than the input supply. This allows the driver

to saturate the internal bipolar NPN power switch for

efﬁ cient operation.

OPERATION

LT3475/LT3475-1

3475fb

Open Circuit Protection

The LT3475 has internal open-circuit protection. If the LED

is absent or is open circuit, the LT3475 clamps the voltage

on the LED pin at 14V. The switching regulator then oper-

ates at a very low frequency to limit the input current. The

LT3475-1 has no internal open circuit protection. With the

LT3475-1, be careful not to violate the ABSMAX voltage of

th BOOST pin; if VIN > 25V, external open circuit protection

circuitry (as shown in Figure 1) may be necessary.The

output voltage during an open LED condition is shown in

the Typical Performance Characteristics section.

Undervoltage Lockout

Undervoltage lockout (UVLO) is typically used in situations

where the input supply is current limited, or has high source

resistance. A switching regulator draws constant power

from the source, so the source current increases as the

source voltage drops. This looks like a negative resistance

load to the source and can cause the source to current limit

or latch low under low source voltage conditions. UVLO

prevents the regulator from operating at source voltages

where these problems might occur.

An internal comparator will force the part into shut-

down when VIN falls below 3.7V. If an adjustable UVLO

threshold is required, the SHDN pin can be used. The

threshold voltage of the SHDN pin comparator is 2.6V. An

internal resistor pulls 9μA to ground from the SHDN pin

at the UVLO threshold.

Choose resistors according to the following formula:

R2 =2.6V

VTH –2.6V

R1 –9μA

VTH = UVLO Threshold

Example: Switching should not start until the input is

above 8V.

VTH = 8V

R1=100k

R2 =2.6V

8V – 2.6V

100k –9μA

=57.6k

Figure 2. Undervoltage Lockout

GND

9μA

2.6V

VIN

LT3475

C1 R2

VIN

SHDN

3475 F02

Table 1. Switching Frequencies

SWITCHING FREQUENCY (MHz) RT (kΩ)

2 4.32

1.5 6.81

1.2 9.09

1 11.8

0.8 16.9

0.6 24.3

0.4 40.2

0.3 57.6

0.2 100

Keep the connections from the resistors to the SHDN pin

short and make sure the coupling to the SW and BOOST

pins is minimized. If high resistance values are used, the

SHDN pin should be bypassed with a 1nF capacitor to

prevent coupling problems from switching nodes.

Setting the Switching Frequency

The LT3475 uses a constant frequency architecture that

can be programmed over a 200kHz to 2MHz range with a

single external timing resistor from the RT pin to ground.

A graph for selecting the value of RT for a given operating

frequency is shown in the Typical Applications section.

100k

10k

22V

OUT

3475 F01

Figure 1. External Overvoltage Protection

Circuitry for the LT3475-1

APPLICATIONS INFORMATION

LT3475/LT3475-1

3475fb

Table 1 shows suggested RT selections for a variety of

switching frequencies.

Operating Frequency Selection

The choice of operating frequency is determined by

several factors. There is a tradeoff between efﬁ ciency and

component size. A higher switching frequency allows the

use of smaller inductors at the cost of increased switching

losses and decreased efﬁ ciency.

Another consideration is the maximum duty cycle. In certain

applications, the converter needs to operate at a high duty

cycle in order to work at the lowest input voltage possible.

The LT3475 has a ﬁ xed oscillator off time and a variable

on time. As a result, the maximum duty cycle increases

as the switching frequency is decreased.

Input Voltage Range

The minimum operating voltage is determined either by the

LT3475’s undervoltage lockout of 4V, or by its maximum

duty cycle. The duty cycle is the fraction of time that the

internal switch is on and is determined by the input and

output voltages:

DC =VOUT +VF

()

VIN –V

SW +VF

()

where VF is the forward voltage drop of the catch diode

(~0.4V) and VSW is the voltage drop of the internal switch

(~0.4V at maximum load). This leads to a minimum input

voltage of:

IN MIN

()

=VOUT +V

DCMAX

–V

F+VSW

with DCMAX = 1–tOFF(MIN) • f

where t0FF(MIN) is equal to 167ns and f is the switching

frequency.

Example: f = 600kHz, VOUT = 4V

DCMAX =1−167ns • 600kHz =0.90

IN MIN

()

=4V +0.4V

0.9 – 0.4V +0.4V =4.9V

The maximum operating voltage is determined by the

absolute maximum ratings of the VIN and BOOST pins,

and by the minimum duty cycle.

IN MAX

()

=VOUT +V

DCMIN

–V

F+VSW

with DCMIN = tON(MIN) • f

where tON(MIN) is equal to 140ns and f is the switching

frequency.

Example: f = 750kHz, VOUT = 3.4V

DCMIN =140ns • 750kHz =0.105

IN MAX

()

=3.4V +0.4V

0.105 – 0.4V +0.4V =36V

The minimum duty cycle depends on the switching fre-

quency. Running at a lower switching frequency might

allow a higher maximum operating voltage. Note that

this is a restriction on the operating input voltage; the

circuit will tolerate transient inputs up to the Absolute

Maximum Ratings of the VIN and BOOST pins. The input

voltage should be limited to the VIN operating range (36V)

during overload conditions (short circuit or start up).

Minimum On Time

The LT3475 will regulate the output current at input volt-

ages greater than VIN(MAX). For example, an application

with an output voltage of 3V and switching frequency of

1.2MHz has a VIN(MAX) of 20V, as shown in Figure 3. Figure

4 shows operation at 35V. Output ripple and peak inductor

VSW

20V/DIV

1A/DIV

VOUT

500mV/DIV

(AC COUPLED)

3475 F03

Figure 3. Operation at VIN(MAX) = 20V.

VOUT = 3V and fSW = 1.2MHHz

APPLICATIONS INFORMATION

LT3475/LT3475-1

3475fb

current have signiﬁ cantly increased. Exceeding VIN(MAX)

is safe if the external components have adequate ratings

to handle the peak conditions and if the peak inductor

current does not exceed 3.2A. A saturating inductor may

further reduce performance.

Table 2. Inductors

PART NUMBER

VALUE

(μH)

IRMS

(A)

DCR

()

HEIGHT

(mm)

Sumida

CR43-3R3 3.3 1.44 0.086 3.5

CR43-4R7 4.7 1.15 0.109 3.5

CDRH4D16-3R3 3.3 1.10 0.063 1.8

CDRH4D28-3R3 3.3 1.57 0.049 3.0

CDRH4D28-4R7 4.7 1.32 0.072 3.0

CDRH6D26-5R0 5.0 2.20 0.032 2.8

CDRH6D26-5R6 5.6 2.0 0.036 2.8

CDRH5D28-100 10 1.30 0.048 3.0

CDRH5D28-150 15 1.10 0.076 3.0

CDRH73-100 10 1.68 0.072 3.4

CDRH73-150 15 1.33 0.130 3.4

CDRH104R-150 15 3.1 0.050 4.0

Coilcraft

DO1606T-332 3.3 1.30 0.100 2.0

DO1606T-472 4.7 1.10 0.120 2.0

DO1608C-332 3.3 2.00 0.080 2.9

DO1608C-472 4.7 1.50 0.090 2.9

MOS6020-332 3.3 1.80 0.046 2.0

MOS6020-472 10 1.50 0.050 2.0

DO3316P-103 10 3.9 0.038 5.2

DO3316P-153 15 3.1 0.046 5.2

The optimum inductor for a given application may differ

from the one indicated by this simple design guide. A larger

value inductor provides a higher maximum load current, and

reduces the output voltage ripple. If your load is lower than

the maximum load current, then you can relax the value of the

inductor and operate with higher ripple current. This allows

you to use a physically smaller inductor, or one with a lower

DCR resulting in higher efﬁ ciency. In addition, low inductance

may result in discontinuous mode operation, which further

reduces maximum load current. For details of maximum

output current and discontinuous mode operation, see Linear

Technology’s Application Note 44. Finally, for duty cycles

greater than 50% (VOUT/VIN > 0.5), a minimum inductance

is required to avoid sub-harmonic oscillations:

LMIN =(VOUT +VF)•800kHz

VSW

20V/DIV

1A/DIV

VOUT

500mV/DIV

(AC COUPLED)

3475 F04

Figure 4. Operation above VIN(MAX). Output

Ripple and Peak Inductor Current Increases

Inductor Selection and Maximum Output Current

A good ﬁ rst choice for the inductor value is:

L=(VOUT +VF)•1.2MHz

where VF is the voltage drop of the catch diode (~0.4V),

f is the switching frequency and L is in μH. With this value

the maximum load current will be above 1.6A at all duty

cycles. The inductor’s RMS current rating must be greater

than the maximum load current and its saturation current

should be at least 30% higher. For highest efﬁ ciency,

the series resistance (DCR) should be less than 0.15Ω.

Table 2 lists several vendors and types that are suitable.

For robust operation at full load and high input voltages

(VIN > 30V), use an inductor with a saturation current

higher than 3.2A.

APPLICATIONS INFORMATION

LT3475/LT3475-1

3475fb

The current in the inductor is a triangle wave with an average

value equal to the load current. The peak switch current

is equal to the output current plus half the peak-to-peak

inductor ripple current. The LT3475 limits its switch cur-

rent in order to protect itself and the system from overload

faults. Therefore, the maximum output current that the

LT3475 will deliver depends on the switch current limit,

the inductor value, and the input and output voltages.

When the switch is off, the potential across the inductor

is the output voltage plus the catch diode drop. This gives

the peak-to-peak ripple current in the inductor

ΔIL=1–DC

()

VOUT +VF

()

L•f

()

where f is the switching frequency of the LT3475 and L

is the value of the inductor. The peak inductor and switch

current is

ISW PK

()

=ILPK

()

=IOUT +ΔIL

To maintain output regulation, this peak current must be

less than the LT3475’s switch current limit ILIM. ILIM is at

least 2.3A at low duty cycles and decreases linearly to 1.8A

at DC = 0.9. The maximum output current is a function of

the chosen inductor value:

IOUT MAX

()

= ILIM –ΔIL

=2.3A• 1–0.25•DC

()

–ΔIL

Choosing an inductor value so that the ripple current is

small will allow a maximum output current near the switch

current limit.

One approach to choosing the inductor is to start with the

simple rule given above, look at the available inductors,

and choose one to meet cost or space goals. Then use

these equations to check that the LT3475 will be able to

deliver the required output current. Note again that these

equations assume that the inductor current is continu-

ous. Discontinuous operation occurs when IOUT is less

than ΔIL/2.

Input Capacitor Selection

Bypass the input of the LT3475 circuit with a 4.7μF or

higher ceramic capacitor of X7R or X5R type. A lower

value or a less expensive Y5V type will work if there is

additional bypassing provided by bulk electrolytic capaci-

tors or if the input source impedance is low. The following

paragraphs describe the input capacitor considerations in

more detail.

Step-down regulators draw current from the input supply

in pulses with very fast rise and fall times. The input ca-

pacitor is required to reduce the resulting voltage ripple at

the LT3475 input and to force this switching current into a

tight local loop, minimizing EMI. The input capacitor must

have low impedance at the switching frequency to do this

effectively, and it must have an adequate ripple current rat-

ing. With two switchers operating at the same frequency

but with different phases and duty cycles, calculating the

input capacitor RMS current is not simple. However, a

conservative value is the RMS input current for the channel

that is delivering most power (VOUT • IOUT):

CINRMS =IOUT •VOUT(V

IN –V

OUT)

VIN

<IOUT

and is largest when VIN = 2VOUT (50% duty cycle). As the

second, lower power channel draws input current, the

input capacitor’s RMS current actually decreases as the

out-of-phase current cancels the current drawn by the

higher power channel. Considering that the maximum

load current from a single channel is ~1.5A, RMS ripple

current will always be less than 0.75A.

The high frequency of the LT3475 reduces the energy

storage requirements of the input capacitor, so that the

capacitance required is less than 10μF. The combination

of small size and low impedance (low equivalent series

resistance or ESR) of ceramic capacitors makes them the

preferred choice. The low ESR results in very low voltage

ripple. Ceramic capacitors can handle larger magnitudes

of ripple current than other capacitor types of the same

value. Use X5R and X7R types.

APPLICATIONS INFORMATION

LT3475/LT3475-1

3475fb

An alternative to a high value ceramic capacitor is a

lower value ceramic along with a larger electrolytic

capacitor. The electrolytic capacitor likely needs to be greater

than 10μF in order to meet the ESR and ripple current

requirements. The input capacitor is likely to see high

surge currents when the input source is applied. Tanta-

lum capacitors can fail due to an over-surge of current.

Only use tantalum capacitors with the appropriate surge

current rating. The manufacturer may also recommend

operation below the rated voltage of the capacitor.

A ﬁ nal caution is in order regarding the use of ceramic

capacitors at the input. A ceramic input capacitor can

combine with stray inductance to form a resonant tank

circuit. If power is applied quickly (for example by plug-

ging the circuit into a live power source) this tank can ring,

doubling the input voltage and damaging the LT3475. The

solution is to either clamp the input voltage or dampen the

tank circuit by adding a lossy capacitor in parallel with the

ceramic capacitor. For details, see Application Note 88.

Output Capacitor Selection

For most LEDs, a 2.2μF, 6.3V ceramic capacitor (X5R or

X7R) at the output results in very low output voltage ripple

and good transient response. Other types and values will

also work. The following discusses tradeoffs in output

ripple and transient performance.

The output capacitor ﬁ lters the inductor current to

generate an output with low voltage ripple. It also stores

energy in order to satisfy transient loads and stabilizes the

LT3475’s control loop. Because the LT3475 operates at a

high frequency, minimal output capacitance is necessary.

In addition, the control loop operates well with or without

the presence of output capacitor series resistance (ESR).

Ceramic capacitors, which achieve very low output ripple

and small circuit size, are therefore an option.

You can estimate output ripple with the following

equation:

RIPPLE = ΔIL / (8 • f • COUT) for ceramic capacitors

where ΔIL is the peak-to-peak ripple current in the

inductor. The RMS content of this ripple is very low so the

RMS current rating of the output capacitor is usually not

of concern. It can be estimated with the formula:

IC(RMS) =ΔIL/12

The low ESR and small size of ceramic capacitors make

them the preferred type for LT3475 applications. Not all

ceramic capacitors are the same, however. Many of the

higher value capacitors use poor dielectrics with high

temperature and voltage coefﬁ cients. In particular Y5V

and Z5U types lose a large fraction of their capacitance

with applied voltage and at temperature extremes.

Because loop stability and transient response depend on

the value of COUT

, this loss may be unacceptable. Use X7R

and X5R types. Table 3 lists several capacitor vendors.

Table 3. Low ESR Surface Mount Capacitors.

VENDOR TYPE SERIES

Taiyo-Yuden Ceramic X5R, X7R

AVX Ceramic X5R, X7R

TDK Ceramic X5R, X7R

Diode Selection

The catch diode (D3 from the Block Diagram) conducts

current only during switch off time. Average forward cur-

rent in normal operation can be calculated from:

D(AVG) = IOUT (VIN – VOUT)/VIN

The only reason to consider a diode with a larger current

rating than necessary for nominal operation is for the

worst-case condition of shorted output. The diode cur-

rent will then increase to one half the typical peak switch

current limit.

Peak reverse voltage is equal to the regulator input

voltage. Use a diode with a reverse voltage rating greater

than the input voltage. Table 4 lists several Schottky

diodes and their manufacturers.

Diode reverse leakage can discharge the output capacitor

during LED off times while PWM dimming. If operating at

high ambient temperatures, use a low leakage Schottky

for the widest PWM dimming range.

APPLICATIONS INFORMATION

LT3475/LT3475-1

3475fb

Table 4. Schottky Diodes

(V)

IAVE(A)

(A)

VF at 1A

(mV)

VF at 2A

(mV)

On Semiconductor

MBR0540 40 0.5 620

MBRM120E 20 1 530

MBRM140 40 1 550

Diodes Inc

B120 20 1 500

B130 30 1 500

B140HB 40 1 530

DFLS140 40 1.1 510

B240 40 2 500

International Rectiﬁ er

10BQ030 30 1 420

BOOST Pin Considerations

The capacitor and diode tied to the BOOST pin gener-

ate a voltage that is higher than the input voltage. In

most cases, a 0.22μF capacitor and fast switching diode

(such as the CMDSH-3 or MMSD914LT1) will work well.

Figure 5 shows three ways to arrange the boost circuit.

The BOOST pin must be more than 2.5V above the SW

pin for full efﬁ ciency. For outputs of 3.3V and higher, the

standard circuit (Figure 5a) is best. For outputs between

2.8V and 3.3V, use a small Schottky diode (such as the

BAT-54). For lower output voltages, the boost diode can be

tied to the input (Figure 5b). The circuit in Figure 5a is more

efﬁ cient because the BOOST pin current comes from a

lower voltage source. The anode of the boost diode can

be tied to another source that is at least 3V. For example, if

you are generating a 3.3V output, and the 3.3V output is on

whenever the LED is on, the BOOST pin can be

connected to the 3.3V output. For LT3475-1 applications

with higher output voltages, an additional Zener diode

may be necessary (Figure 5d) to maintain pin voltage

below the absolute maximum. In any case, be sure that

the maximum voltage at the BOOST pin is both less than

60V and the voltage difference between the BOOST and

SW pins is less than 30V.

The minimum operating voltage of an LT3475 application

is limited by the undervoltage lockout (~3.7V) and by the

maximum duty cycle. The boost circuit also limits the

minimum input voltage for proper start up. If the input

voltage ramps slowly, or the LT3475 turns on when the

output is already in regulation, the boost capacitor may

not be fully charged. Because the boost capacitor charges

Figure 5. Generating the Boost Voltage

VIN

BOOST

GND

VIN

LT3475

(5a)

VOUT

VBOOST – VSW ≅ VOUT

MAX VBOOST ≅ VIN + VOUT

VIN

BOOST

GND

VIN

LT3475

(5b)

VOUT

VBOOST – VSW ≅ VIN

MAX VBOOST ≅ 2VIN

VIN

BOOST

GND

VIN

LT3475

(5c)

3475 F05

VOUT

VBOOST – VSW ≅ VIN2

MAX VBOOST ≅ VIN2 + VIN

MINIMUM VALUE FOR VIN2 = 3V

VIN2 > 3V

VIN

BOOST

GND

VIN

LT3475

(5d)

3475 F05

VOUT

VBOOST – VSW – VZ

MAX VBOOST ≅ VIN + VOUT – VZ

APPLICATIONS INFORMATION

LT3475/LT3475-1

3475fb

with the energy stored in the inductor, the circuit will rely

on some minimum load current to get the boost circuit

running properly. This minimum load will depend on input

and output voltages, and on the arrangement of the boost

circuit. The minimum load current generally goes to zero

once the circuit has started. The typical performance char-

acteristics section shows a plot of minimum load to start

and to run as a function of input voltage. Even without an

output load current, in many cases the discharged output

capacitor will present a load to the switcher that will allow

it to start. The plots show the worst case, where VIN is

ramping very slowly.

Programming LED Current

The LED current can be set by adjusting the voltage on the

VADJ pin. For a 1.5A LED current, either tie VADJ to REF or

to a 1.25V source. For lower output currents, program the

VADJ using the following formula: ILED = 1.5A • VADJ/1.25V.

Voltages less than 1.25V can be generated with a voltage

divider from the REF pin, as shown in Figure 6. In order

to have accurate LED current, precision resistors are

preferred (1% or better is recommended). Note that the

VADJ pin sources a small amount of bias current, so use

the following formula to choose resistors:

R2 =VADJ

1.25V – VADJ

R1 +50nA

To minimize the error from variations in VADJ pin current,

use resistors with a parallel resistance of less than 4k. Use

resistor strings with a high enough series resistance so as not

to exceed the 500μA current compliance of the REF pin.

Dimming Control

There are several different types of dimming control

circuits. One dimming control circuit (Figure 7) changes

the voltage on the VADJ pin by tying a low on resistance

FET to the resistor divider string. This allows the se-

lection of two different LED currents. For reliable op-

eration program an LED current of no less than 50mA.

The maximum current dimming ratio (IRATIO) can be

calculated from the maximum LED current (IMAX) and the

minimum LED current (IMIN) as follows:

MAX/IMIN = IRATIO

Another dimming control circuit (Figure 8) uses the PWM

pin and an external NFET tied to the cathode of the LED.

An external PWM signal is applied to the PWM pin and the

gate of the NFET (For PWM dimming ratios of 20 to 1 or

less, the NFET can be omitted). The average LED current is

proportional to the duty cycle of the PWM signal. When the

PWM signal goes low, the NFET turns off, turning off the

LED and leaving the output capacitor charged. The PWM

pin is pulled low as well, which disconnects the VC pin,

storing the voltage in the capacitor tied there. Use the C-RC

string shown in Figure 8 and Figure 9 tied to the VC pin for

proper operation during startup. When the PWM pin goes

high again, the LED current returns rapidly to its previous

on state since the compensation and output capacitors are

at the correct voltage. This fast settling time allows the

PWM

LED

GND

LT3475

3475 F08

PWM

100Hz TO

10kHz

10k

0.1μF

3.3nF

Figure 8. Dimming Using PWM Signal

REF

VADJ

GND

LT3475

3475 F07

DIM

Figure 7. Dimming with a MOSFET and Resistor Divider

Figure 6. Setting VADJ with a Resistor Divider

REF

VADJ

GND

LT3475

3475 F06

APPLICATIONS INFORMATION

LT3475/LT3475-1

3475fb

Figure 10. Recommended Component Placement

3475 F10

VIN

PWM2SHDNPWM1

VIA TO LOCAL GND PLANE

RTVC

GND

LT3475

3475 F09

10k

220pF

0.1μ

PWM1

3.3nF

Figure 9. Extending the PWM Dimming Range

LT3475 to maintain diode current regulation with PWM

pulse widths as short as 7.5 switching cycles (12.5μs for

fSW = 600kHz). Maximum PWM period is determined by

the system and is unlikely to be longer than 12ms. Using

PWM periods shorter than 100μs is not recommended.

The maximum PWM dimming ratio (PWMRATIO) can be

calculated from the maximum PWM period (tMAX) and

minimum PWM pulse width (tMIN) as follows:

MAX/tMIN = PWMRATIO

Total dimming ratio (DIMRATIO) is the product of the PWM

dimming ratio and the current dimming ratio.

Example:

MAX = 1A, IMIN = 0.1A, tMAX = 9.9ms

MIN = 3.3μs (fSW = 1.4MHz)

RATIO = 1A/0.1A =10:1

PWMRATIO = 9.9ms/3.3μs = 3000:1

DIMRATIO = 10 • 3000 = 30000:1

To achieve the maximum PWM dimming ratio, use the

circuit shown in Figure 9. This allows PWM pulse widths

as short as 4.5 switching cycles (7.5μs for fSW = 600kHz).

Note that if you use the circuit in Figure 9, the rising edge

of the two PWM signals must align within 100ns.

Layout Hints

As with all switching regulators, careful attention must

be paid to the PCB layout and component placement. To

maximize efﬁ ciency, switch rise and fall times are made

as short as possible. To prevent electromagnetic interfer-

ence (EMI) problems, proper layout of the high frequency

switching path is essential. The voltage signal of the SW

and BOOST pins have sharp rise and fall edges. Minimize

the area of all traces connected to the BOOST and SW

pins and always use a ground plane under the switching

regulator to minimize interplane coupling. In addition, the

ground connection for frequency setting resistor RT and

capacitors at VC1, VC2 pins (refer to the Block Diagram)

should be tied directly to the GND pin and not shared

with the power ground path, ensuring a clean, noise-free

connection.

APPLICATIONS INFORMATION

LT3475/LT3475-1

3475fb

BOOST1 BOOST2

SW1

LED 1 LED 2

D3 D4

SW2

OUT1 OUT2

LED1 LED2

VC1 VC2

REF RT

VADJ1 VADJ2

GND

LT3475

VIN SHDN

10μH

0.22μF

6.3V

0.22μF

6.3V

24.3k

2.2μF

6.3V

2.2μF

6.3V

0.1μF

3475 TA02

C1 TO C5: X5R OR X7R

D1, D2: DFLS140

D3, D4: MBR0540

LED CURRENT = 1A

VIN

5V TO 36V C1

4.7μF

50V

fSW = 600kHz

BOOST1 BOOST2

SW1

LED 11.5A LED

CURRENT

1.5A LED

CURRENT

LED 2

D3 D4

SW2

OUT1 OUT2

LED1 LED2

PWM1 PWM2

VC1 VC2

REF RT

VADJ1 VADJ2

GND

LT3475

VIN SHDN

10μH

0.22μF

6.3V

0.22μF

6.3V

2.2μF

6.3V

2.2μF

6.3V

3.3nF

0.1μF

220p

3.3nF

3475 TA03

D1, D2: B260

D3, D4: MBR0540

C1 TO C5: X5R OR X7R

M1, M2: Si2302ADS

M3: 2n7002L

VIN

6V TO 36V C1

4.7μF

50V

M1 M2

M3 R1

24.3k

10k

PWM1 PWM2

fSW = 600kHz

Dual Step-Down 1A LED Driver

Dual Step-Down 1.5A LED Driver with 1200 : 1 True Color PWM Dimming

TYPICAL APPLICATIONS

LT3475/LT3475-1

3475fb

BOOST1 BOOST2

SW1

D3 D4

SW2

OUT1 OUT2

LED1

LED2

VC1 VC2

REF RT

VADJ1 VADJ2

GND

LT3475

VIN SHDN

10μH

0.22μF

6.3V

0.22μF

6.3V

2.2μF

6.3V

2.2μF

6.3V

3475 TA04

D1, D2: B240A

D3, D4: MBR0540

C1 TO C5: X5R OR X7R

VIN

5V TO 36V C1

4.7μF

50V

24.3k

0.1μF

3A LED

CURRENT

LED 1

fSW = 600kHz

BOOST1 BOOST2

SW1

D3 D4

SW2

OUT1

LED1

OUT2

LED2

VC1 VC2

REF RT

VADJ1 VADJ2

GND

LT3475

VIN SHDN

15μH

0.22μF

10V

0.22μF

10V

2.2μF

10V

2.2μF

10V

3475 TA05

D1, D2: B240A

D3, D4: MMSD4148T1

C1 TO C5: X5R OR X7R

VIN

10V TO 36V C1

4.7μF

50V

24.3k

0.1μF

LED 4LED 3

1.5A LED

CURRENT

LED 2

1.5A LED

CURRENT

LED 1

fSW = 600kHz

Step-Down 3A LED Driver

Dual Step-Down LED Driver with Series Connected LEDs

TYPICAL APPLICATIONS

LT3475/LT3475-1

3475fb

BOOST1 BOOST2

SW1

SW2

OUT1

LED1

OUT2

LED2

VC1 VC2

REF RT

VADJ1 VADJ2

GND

LT3475

VIN SHDN

10μH

0.22μF

35V

0.22μF

35V

2.2μF

6.3V

2.2μF

6.3V

3475 TA06

D1, D2: B240A

D3, D4: MMSD4148T1

C1 TO C5: X5R OR X7R

VIN

5V TO 28V C1

4.7μF

35V

24.3k

0.1μF

LED 2LED 1 1.5A LED

CURRENT

1.5A LED

CURRENT

fSW = 600kHz

Dual Step-Down 1.5A Red LED Driver

TYPICAL APPLICATIONS

LT3475/LT3475-1

3475fb

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However,

no responsibility is assumed for its use. Linear Technology Corporation makes no representation that

the interconnection of its circuits as described herein will not infringe on existing patent rights.

FE20 (CB) TSSOP 0204

0.09 – 0.20

(.0035 – .0079)

0° – 8°

0.25

REF

RECOMMENDED SOLDER PAD LAYOUT

0.50 – 0.75

(.020 – .030)

4.30 – 4.50*

(.169 – .177)

134

5678910

111214 13

6.40 – 6.60*

(.252 – .260)

3.86

(.152)

2.74

(.108)

20 1918 17 16 15

1.20

(.047)

MAX

0.05 – 0.15

(.002 – .006)

0.65

(.0256)

BSC 0.195 – 0.30

(.0077 – .0118)

TYP

2.74

(.108)

0.45 ±0.05

0.65 BSC

4.50 ±0.10

6.60 ±0.10

1.05 ±0.10

3.86

(.152)

MILLIMETERS

(INCHES) *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH

SHALL NOT EXCEED 0.150mm (.006") PER SIDE

NOTE:

1. CONTROLLING DIMENSION: MILLIMETERS

2. DIMENSIONS ARE IN

3. DRAWING NOT TO SCALE

SEE NOTE 4

4. RECOMMENDED MINIMUM PCB METAL SIZE

FOR EXPOSED PAD ATTACHMENT

6.40

(.252)

BSC

FE Package

20-Lead Plastic TSSOP (4.4mm)

(Reference LTC DWG # 05-08-1663)

Exposed Pad Variation CB

PACKAGE DESCRIPTION

LT3475/LT3475-1

3475fb

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com

LT 1007 REV B • PRINTED IN USA

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BOOST1 BOOST2

SW1 SW2

OUT1 OUT2

LED1

12V TO 18V LED VOLTAGE 12V TO 18V LED VOLTAGE

LED2

VC1 VC2

REF RT

VADJ1 VADJ2

GND

LT3475-1

VIN

21V TO 36V

SHDN

0.22μF

16V

4.7μF

50V

33μH

100k

10k

100k

10k

33μH

0.22μF

16V

24.3k

2.2μF

25V

2.2μF

25V

D2D1

0.1μF

3475 TA08

1.5A LED

CURRENT*

1.5A LED

CURRENT*

fSW = 600kHz

D1, D4: 7.5V ZENER DIODE

D2, D3: MMSD4148

D5, D6: B240A

D7, D8: 22V ZENER DIODE

R1, R2: USE 0.5W RESISTOR OF TWO 2k 0.25W RESISTORS IN PARALLEL

Q1, Q2: MMBT3904

C1 TO C5: X5R or X7R

*DERATE LED CURRENT AT ELEVATED AMBIENT TEMPERATURES TO MAINTAIN LT3475-1 JUNCTION TEMPERATURE BELOW 125 °C.

Dual Step-Down 1.5A LED Driver with Four Series Connected LED Output

TYPICAL APPLICATION