LM3886
Overture
™Audio Power Amplifier Series
High-Performance 68W Audio Power Amplifier w/Mute
General Description
The LM3886 is a high-performance audio power amplifier
capable of delivering 68W of continuous average power to a
4Ωload and 38W into 8Ωwith 0.1%(THD + N) from
20 Hz–20 kHz.
The performance of the LM3886, utilizing its Self Peak In-
stantaneous Temperature (˚Ke) (SPiKe™) Protection Cir-
cuitry, puts it in a class above discrete and hybrid amplifiers
by providing an inherently, dynamically protected Safe Oper-
ating Area (SOA). SPiKe Protection means that these parts
are completely safeguarded at the output against overvolt-
age, undervoltage, overloads, including shorts to the sup-
plies, thermal runaway, and instantaneous temperature
peaks.
The LM3886 maintains an excellent Signal-to-Noise Ratio of
greater than 92 dB with a typical low noise floor of 2.0 µV. It
exhibits extremely low (THD + N) values of 0.03%at the
rated output into the rated load over the audio spectrum, and
provides excellent linearity with an IMD (SMPTE) typical rat-
ing of 0.004%.
Features
n68W cont. avg. output power into 4Ωat V
CC
=±28V
n38W cont. avg. output power into 8Ωat V
CC
=±28V
n50W cont. avg. output power into 8Ωat V
CC
=±35V
n135W instantaneous peak output power capability
nSignal-to-Noise Ratio ≥92 dB
nAn input mute function
nOutput protection from a short to ground or to the
supplies via internal current limiting circuitry
nOutput over-voltage protection against transients from
inductive loads
nSupply under-voltage protection, not allowing internal
biasing to occur when |V
EE
|+|V
CC
|≤12V, thus
eliminating turn-on and turn-off transients
n11-lead TO-220 package
Applications
nComponent stereo
nCompact stereo
nSelf-powered speakers
nSurround-sound amplifiers
nHigh-end stereo TVs
Typical Application
Overture™and SPiKe ™Protection are trademarks of National Semiconductor Corporation.
DS011833-1
*Optional components dependent upon specific design requirements. Refer to the External Components Description section for a component functional
description.
FIGURE 1. Typical Audio Amplifier Application Circuit
May 1999
LM3886
Overture
Audio Power Amplifier Series High-Performance 68W Audio Power Amplifier
w/Mute
© 1999 National Semiconductor Corporation DS011833 www.national.com
Connection Diagram
Plastic Package (Note 12)
DS011833-2
Note 1: Preliminary: call you local National Sales Rep. or distributor for availability
Top View
Order Number LM3886T
or LM3886TF
See NS Package Number TA11B for
Staggered Lead Non-Isolated
Package or TF11B (Note 1) for
Staggered Lead Isolated Package
www.national.com 2
Absolute Maximum Ratings (Notes 6,
5)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Supply Voltage |V
+
|+|V
−
|
(No Signal) 94V
Supply Voltage |V
+
|+|V
−
|
(Input Signal) 84V
Common Mode Input Voltage (V
+
or V
−
) and
|V
+
|+|V
−
|≤80V
Differential Input Voltage (Note 16) 60V
Output Current Internally Limited
Power Dissipation (Note 7) 125W
ESD Susceptibility (Note 8) 3000V
Junction Temperature (Note 9) 150˚C
Soldering Information
T Package (10 seconds) 260˚C
Storage Temperature −40˚C to +150˚C
Thermal Resistance
θ
JC
1˚C/W
θ
JA
43˚C/W
Operating Ratings (Notes 5, 6)
Temperature Range
T
MIN
≤T
A
≤T
MAX
−20˚C ≤T
A
≤+85˚C
Supply Voltage |V
+
|+|V
−
| 20V to 84V
Electrical Characteristics (Notes 5, 6)
The following specifications apply for V
+
=+28V, V
−
=−28V, I
MUTE
=−0.5 mA with R
L
=4Ωunless otherwise specified. Limits
apply for T
A
=25˚C.
Symbol Parameter Conditions LM3886 Units
(Limits)
Typical
(Note 10) Limit
(Note 11)
|V
+
|+|V
−
| Power Supply Voltage (Note 14) V
pin7
−V
−
≥9V 18 20
84 V (min)
V (max)
A
M
Mute Attenuation Pin 8 Open or at 0V, Mute: On
Current out of Pin 8 >0.5 mA,
Mute: Off 115 80 dB (min)
P
O
(Note 4) Output Power (Continuous Average) THD + N =0.1%(max)
f=1 kHz; f =20 kHz
|V
+
|=|V
−
|=28V, R
L
=4Ω
|V
+
|=|V
−
|=28V, R
L
=8Ω
|V
+
|=|V
−
|=35V, R
L
=8Ω
68
38
50
60
30 W (min)
W (min)
W
Peak P
O
Instantaneous Peak Output Power 135 W
THD + N Total Harmonic Distortion Plus Noise 60W, R
L
=4Ω,
30W, R
L
=8Ω,
20 Hz ≤f≤20 kHz
A
V
=26 dB
0.03
0.03
%
%
SR
(Note 4) Slew Rate (Note 13) V
IN
=2.0Vp-p, t
RISE
=2ns 19 8 V/µs
(min)
I
+
(Note 4) Total Quiescent Power Supply
Current V
CM
=0V, V
o
=0V, I
o
=0A 50 85 mA (max)
V
OS
(Note 3) Input Offset Voltage V
CM
=0V, I
o
=0mA 1 10 mV (max)
I
B
Input Bias Current V
CM
=0V, I
o
=0 mA 0.2 1 µA (max)
I
OS
Input Offset Current V
CM
=0V, I
o
=0 mA 0.01 0.2 µA (max)
I
o
Output Current Limit |V
+
|=|V
−
|=20V, t
ON
=10 ms, V
O
=0V 11.5 7 A (min)
V
od
(Note 3) Output Dropout Voltage (Note 15) |V
+
–V
O
|, V
+
=28V, I
o
=+100 mA
|V
O
–V
−
|, V
−
=−28V, I
o
=−100 mA 1.6
2.5 2.0
3.0 V (max)
V (max)
PSRR
(Note 3) Power Supply Rejection Ratio V
+
=40V to 20V, V
−
=−40V,
V
CM
=0V, I
o
=0mA
V
+
=40V, V
−
=−40V to −20V,
V
CM
=0V, I
o
=0mA
120
105
85
85
dB (min)
dB (min)
CMRR
(Note 3) Common Mode Rejection Ratio V
+
=60V to 20V, V
−
=−20V to −60V,
V
CM
=20V to −20V, I
o
=0mA 110 85 dB (min)
A
VOL
(Note 3) Open Loop Voltage Gain |V
+
|=|V
−
|=28V, R
L
=2kΩ,∆V
O
=40V 115 90 dB (min)
GBWP Gain-Bandwidth Product |V
+
|=|V
−
|=30V
f
O
=100 kHz, V
IN
=50 mVrms 82
MHz
(min)
www.national.com3
Electrical Characteristics (Notes 5, 6) (Continued)
The following specifications apply for V
+
=+28V, V
−
=−28V, I
MUTE
=−0.5 mA with R
L
=4Ωunless otherwise specified. Limits
apply for T
A
=25˚C.
Symbol Parameter Conditions LM3886 Units
(Limits)
Typical
(Note 10) Limit
(Note 11)
e
IN
(Note 4) Input Noise IHF—A Weighting Filter
R
IN
=600Ω(Input Referred) 2.0 10 µV (max)
SNR Signal-to-Noise Ratio P
O
=1W, A-Weighted,
Measured at 1 kHz, R
S
=25Ω92.5 dB
P
O
=60W, A-Weighted,
Measured at 1 kHz, R
S
=25Ω110 dB
IMD Intermodulation Distortion Test 60 Hz, 7 kHz, 4:1 (SMPTE)
60 Hz, 7 kHz, 1:1 (SMPTE) 0.004
0.009 %
Note 2: Operation is guaranteed up to 84V, however, distortion may be introduced from SPIKeProtection Circuitry if proper thermal considerations are not taken into
account. Refer to the Thermal Considerations section for more information. (See SPIKe Protection Response)
Note 3: DC Electrical Test; refer to Test Circuit #1.
Note 4: AC Electrical Test; refer to Test Circuit #2.
Note 5: All voltages are measured with respect to the GND pin (pin 7), unless otherwise specified.
Note 6: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions which
guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters where no limit
is given, however, the typical value is a good indication of device performance.
Note 7: For operating at case temperatures above 25˚C, the device must be derated based on a 150˚C maximum junction temperature and a thermal resistance of
θJC =1.0 ˚C/W (junction to case). Refer to the Thermal Resistance figure in the Application Information section under Thermal Considerations.
Note 8: Human body model, 100 pF discharged through a 1.5 kΩresistor.
Note 9: The operating junction temperature maximum is 150˚C, however, the instantaneous Safe Operating Area temperature is 250˚C.
Note 10: Typicals are measured at 25˚C and represent the parametric norm.
Note 11: Limits are guaranteed to National’s AOQL (Average Outgoing Quality Level).
Note 12: The LM3886T package TA11B is a non-isolated package, setting the tab of the device and the heat sink at V−potential when the LM3886 is directly
mounted to the heat sink using only thermal compound. If a mica washer is used in addition to thermal compound, θCS (case to sink) is increased, but the heat sink
will be isolated from V−.
Note 13: The feedback compensation network limits the bandwidth of the closed-loop response and so the slew rate will be reduced due to the high frequency
roll-off. Without feedback compensation, the slew rate is typically larger.
Note 14: V−must have at least −9V at its pin with reference to ground in order for the under-voltage protection circuitry to be disabled.
Note 15: The output dropout voltage is the supply voltage minus the clipping voltage. Refer to the Clipping Voltage vs Supply Voltage graph in the Typical Perfor-
mance Characteristics section.
Note 16: The Differential Input Voltage Absolute Maximum Rating is based on supply voltages of V+=+40V and V−=−40V.
Test Circuit #1(DC Electrical Test Circuit)
DS011833-3
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Test Circuit #2(AC Electrical Test Circuit)
Single Supply Application Circuit
DS011833-4
DS011833-5
*Optional components dependent upon specific design requirements. Refer to the External
Components Description section for a component functional description.
FIGURE 2. Typical Single Supply Audio Amplifier Application Circuit
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Equivalent Schematic (excluding active protection circuitry)
DS011833-6
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External Components Description
(
Figure 1
and
Figure 2
)
Components Functional Description
1. R
IN
Acts as a volume control by setting the voltage level allowed to the amplifier’s input terminals.
2. R
A
Provides DC voltage biasing for the single supply operation and bias current for the positive input terminal.
3. C
A
Provides bias filtering.
4. C Provides AC coupling at the input and output of the amplifier for single supply operation.
5. R
B
Prevents currents from entering the amplifier’s non-inverting input which may be passed through to the load
upon power-down of the system due to the low input impedance of the circuitry when the under-voltage
circuitry is off. This phenomenon occurs when the supply voltages are below 1.5V.
6. C
C
(Note 17) Reduces the gain (bandwidth of the amplifier) at high frequencies to avoid quasi-saturation oscillations of
the output transistor. The capacitor also suppresses external electromagnetic switching noise created from
fluorescent lamps.
7. Ri Inverting input resistance to provide AC Gain in conjunction with R
f1
.
8. Ci
(Note 17) Feedback capacitor. Ensures unity gain at DC. Also a low frequency pole (highpass roll-off) at:
f
c
=1/(2πRi Ci)
9. R
f1
Feedback resistance to provide AC Gain in conjunction with Ri.
10. R
f2
(Note 17) At higher frequencies feedback resistance works with C
f
to provide lower AC Gain in conjunction with R
f1
and Ri. A high frequency pole (lowpass roll-off) exists at:
f
c
=[R
f1
R
f2
(s + 1/R
f2
C
f
)]/[(R
f1
+R
f2
)(s + 1/C
f
(R
f1
+R
f2
))]
11. C
f
(Note 17) Compensation capacitor that works with R
f1
and R
f2
to reduce the AC Gain at higher frequencies.
12. R
M
Mute resistance set up to allow 0.5 mA to be drawn from pin 8 to turn the muting function off.
→R
M
is calculated using: R
M
≤(|V
EE
| − 2.6V)/I8 where I8 ≥0.5 mA. Refer to the Mute Attenuation vs.
Mute Current curves in the Typical Performance Characteristics section.
13. C
M
Mute capacitance set up to create a large time constant for turn-on and turn-off muting.
14. R
SN
(Note 17) Works with C
SN
to stabilize the output stage by creating a pole that eliminates high frequency oscillations.
15. C
SN
(Note 17) Works with R
SN
to stabilize the output stage by creating a pole that eliminates high frequency oscillations.
f
c
=1/(2πR
SN
C
SN
)
16. L
(Note 17) Provides high impedance at high frequencies so that R may decouple a highly capacitive load and reduce
the Q of the series resonant circuit due to capacitive load. Also provides a low impedance at low
frequencies to short out R and pass audio signals to the load.
17. R
(Note 17)
18. C
S
Provides power supply filtering and bypassing.
19. S1 Mute switch that mutes the music going into the amplifier when opened.
Note 17: Optional components dependent upon specific design requirements. Refer to the Application Information section for more information.
OPTIONAL EXTERNAL COMPONENT INTERACTION
Although the optional external components have specific desired functions that are designed to reduce the bandwidth and elimi-
nate unwanted high frequency oscillations they may cause certain undesirable effects when they interact. Interaction may occur
for components whose reactances are in close proximity to one another. One example would be the coupling capacitor, C
C
, and
the compensation capacitor, C
f
. These two components act as low impedances to certain frequencies which will couple signals
from the input to the output. Please take careful note of basic amplifier component functionality when designing in these compo-
nents.
The optional external components shown in
Figure 2
and described above are applicable in both single and split voltage supply
configurations.
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Typical Performance Characteristics
Safe Area
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SPiKe
Protection Response
DS011833-19
Supply Current vs
Supply Voltage
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Pulse Thermal Resistance
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Pulse Thermal Resistance
DS011833-65
Supply Current vs
Output Voltage
DS011833-22
Pulse Power Limit
DS011833-23
Pulse Power Limit
DS011833-24
Supply Current vs
Case Temperature
DS011833-25
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Typical Performance Characteristics (Continued)
Input Bias Current vs
Case Temperature
DS011833-26
Clipping Voltage vs
Supply Voltage
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Clipping Voltage vs
Supply Voltage
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THD+NvsFrequency
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THD+NvsFrequency
DS011833-30
THD+NvsFrequency
DS011833-31
THD+NvsOutput Power
DS011833-32
THD+NvsOutput Power
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THD+NvsOutput Power
DS011833-34
THD+NvsOutput Power
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THD+NvsOutput Power
DS011833-36
THD+NvsOutput Power
DS011833-37
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Typical Performance Characteristics (Continued)
THD+NvsOutput Power
DS011833-38
THD+NvsOutput Power
DS011833-39
THD+NvsOutput Power
DS011833-40
THD + N Distribution
DS011833-41
THD + N Distribution
DS011833-42
THD + N Distribution
DS011833-43
THD + N Distribution
DS011833-44
THD + N Distribution
DS011833-45
Output Power vs
Load Resistance
DS011833-46
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Typical Performance Characteristics (Continued)
Max Heatsink Thermal Resistance (˚C/W)
at the Specified Ambient Temperature (˚C)
Maximum Power Dissipation vs Supply Voltage
DS011833-9
Note: The maximum heat sink thermal resistance values, øSA, in the table above were calculated using a øCS = 0.2˚C/W due to thermal compound.
Power Dissipation
vs Output Power
DS011833-47
Power Dissipation
vs Output Power
DS011833-48
Output Power
vs Supply Voltage
DS011833-49
IMD 60 Hz, 4:1
DS011833-50
IMD 60 Hz, 7 kHz, 4:1
DS011833-51
IMD 60 Hz, 7 kHz, 4:1
DS011833-52
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Typical Performance Characteristics (Continued)
Application Information
GENERAL FEATURES
Mute Function: The muting function of the LM3886 allows
the user to mute the music going into the amplifier by draw-
ing less than 0.5 mAout of pin 8 of the device. This is accom-
plished as shown in the Typical Application Circuit where the
resistor R
M
is chosen with reference to your negative supply
voltage and is used in conjuction with a switch. The switch
(when opened) cuts off the current flow from pin 8 to V
−
, thus
placing the LM3886 into mute mode. Refer to the Mute At-
tenuation vs Mute Current curves in the Typical Perfor-
mance Characteristics section for values of attenuation per
current out of pin 8. The resistance R
M
is calculated by the
following equation: R
M
(|V
EE
| − 2.6V)/I8
where I8 ≥0.5 mA.
Under-Voltage Protection: Upon system power-up the
under-voltage protection circuitry allows the power supplies
and their corresponding caps to come up close to their full
values before turning on the LM3886 such that no DC output
spikes occur. Upon turn-off, the output of the LM3886 is
brought to ground before the power supplies such that no
transients occur at power-down.
IMD 60 Hz, 1:1
DS011833-53
IMD 60 Hz, 7 kHz 1:1
DS011833-54
IMD 60 Hz, 7 kHz, 1:1
DS011833-55
Mute Attenuation vs
Mute Current
DS011833-56
Mute Attenuation vs
Mute Current
DS011833-57
Large Signal Response
DS011833-58
Power Supply
Rejection Ratio
DS011833-59
Common-Mode
Rejection Ratio
DS011833-60
Open Loop
Frequency Response
DS011833-61
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Application Information (Continued)
Over-Voltage Protection: The LM3886 contains overvolt-
age protection circuitry that limits the output current to ap-
proximately 11Apeak while also providing voltage clamping,
though not through internal clamping diodes. The clamping
effect is quite the same, however, the output transistors are
designed to work alternately by sinking large current spikes.
SPiKe Protection: The LM3886 is protected from instanta-
neous peak-temperature stressing by the power transistor
array. The Safe OperatingArea graph in the Typical Perfor-
mance Characteristics section shows the area of device
operation where the SPiKe Protection Circuitry is not en-
abled. The waveform to the right of the SOA graph exempli-
fies how the dynamic protection will cause waveform distor-
tion when enabled.
Thermal Protection: The LM3886 has a sophisticated ther-
mal protection scheme to prevent long-term thermal stress
to the device. When the temperature on the die reaches
165˚C, the LM3886 shuts down. It starts operating again
when the die temperature drops to about 155˚C, but if the
temperature again begins to rise, shutdown will occur again
at 165˚C. Therefore the device is allowed to heat up to a
relatively high temperature if the fault condition is temporary,
but a sustained fault will cause the device to cycle in a
Schmitt Trigger fashion between the thermal shutdown tem-
perature limits of 165˚C and 155˚C. This greatly reduces the
stress imposed on the IC by thermal cycling, which in turn
improves its reliability under sustained fault conditions.
Since the die temperature is directly dependent upon the
heat sink, the heat sink should be chosen as discussed in
the Thermal Considerations section, such that thermal
shutdown will not be reached during normal operation. Using
the best heat sink possible within the cost and space con-
straints of the system will improve the long-term reliability of
any power semiconductor device.
THERMAL CONSIDERATIONS
Heat Sinking
The choice of a heat sink for a high-power audio amplifier is
made entirely to keep the die temperature at a level such
that the thermal protection circuitry does not operate under
normal circumstances. The heat sink should be chosen to
dissipate the maximum IC power for a given supply voltage
and rated load.
With high-power pulses of longer duration than 100 ms, the
case temperature will heat up drastically without the use of a
heat sink. Therefore the case temperature, as measured at
the center of the package bottom, is entirely dependent on
heat sink design and the mounting of the IC to the heat sink.
For the design of a heat sink for your audio amplifier applica-
tion refer to the Determining The Correct Heat Sink sec-
tion.
Since a semiconductor manufacturer has no control over
which heat sink is used in a particular amplifier design, we
can only inform the system designer of the parameters and
the method needed in the determination of a heat sink. With
this in mind, the system designer must choose his supply
voltages, a rated load, a desired output power level, and
know the ambient temperature surrounding the device.
These parameters are in addition to knowing the maximum
junction temperature and the thermal resistance of the IC,
both of which are provided by National Semiconductor.
As a benefit to the system designer we have provided Maxi-
mum Power Dissipation vs Supply Voltages curves for vari-
ous loads in the Typical Performance Characteristics sec-
tion, giving an accurate figure for the maximum thermal
resistance required for a particular amplifier design. This
data was based on θ
JC
=1˚C/W and θ
CS
=0.2˚C/W. We also
provide a section regarding heat sink determination for any
audio amplifier design where θ
CS
may be a different value. It
should be noted that the idea behind dissipating the maxi-
mum power within the IC is to provide the device with a low
resistance to convection heat transfer such as a heat sink.
Therefore, it is necessary for the system designer to be con-
servative in his heat sink calculations. As a rule, the lower
the thermal resistance of the heat sink the higher the amount
of power that may be dissipated. This is of course guided by
the cost and size requirements of the system. Convection
cooling heat sinks are available commercially, and their
manufacturers should be consulted for ratings.
Proper mounting of the IC is required to minimize the thermal
drop between the package and the heat sink. The heat sink
must also have enough metal under the package to conduct
heat from the center of the package bottom to the fins with-
out excessive temperature drop.
A thermal grease such as Wakefield type 120 or Thermalloy
Thermacote should be used when mounting the package to
the heat sink. Without this compound, thermal resistance will
be no better than 0.5˚C/W, and probably much worse. With
the compound, thermal resistance will be 0.2˚C/W or less,
assuming under 0.005 inch combined flatness runout for the
package and heat sink. Proper torquing of the mounting
bolts is important and can be determined from heat sink
manufacturer’s specification sheets.
Should it be necessary to isolate V
−
from the heat sink, an in-
sulating washer is required. Hard washers like beryluum ox-
ide, anodized aluminum and mica require the use of thermal
compound on both faces. Two-mil mica washers are most
common, giving about 0.4˚C/W interface resistance with the
compound.
Silicone-rubber washers are also available. A 0.5˚C/W ther-
mal resistance is claimed without thermal compound. Expe-
rience has shown that these rubber washers deteriorate and
must be replaced should the IC be dismounted.
Determining Maximum Power Dissipation
Power dissipation within the integrated circuit package is a
very important parameter requiring a thorough understand-
ing if optimum power output is to be obtained. An incorrect
maximum power dissipation (P
D
) calculation may result in in-
adequate heat sinking, causing thermal shutdown circuitry to
operate and limit the output power.
The following equations can be used to acccurately calculate
the maximum and average integrated circuit power dissipa-
tion for your amplifier design, given the supply voltage, rated
load, and output power. These equations can be directly ap-
plied to the Power Dissipation vs Output Power curves in the
Typical Performance Characteristics section.
Equation (1)
exemplifies the maximum power dissipation of
the IC and
Equations (2), (3)
exemplify the average IC power
dissipation expressed in different forms.
P
DMAX
=V
CC
2/2π
2
R
L
(1)
where V
CC
is the total supply voltage
P
DAVE
=(V
Opk
/R
L
)[V
CC
/π−V
Opk
/2] (2)
where V
CC
is the total supply voltage and V
Opk
=V
CC
/π
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Application Information (Continued)
P
DAVE
=V
CC
V
Opk
/πR
L
−V
Opk2
/2R
L
(3)
where V
CC
is the total supply voltage.
Determining the Correct Heat Sink
Once the maximum IC power dissipation is known for a
given supply voltage, rated load, and the desired rated out-
put power the maximum thermal resistance (in ˚C/W) of a
heat sink can be calculated. This calculation is made using
Equation (4)
and is based on the fact that thermal heat flow
parameters are analogous to electrical current flow proper-
ties.
It is also known that typically the thermal resistance, θ
JC
(junction to case), of the LM3886 is 1˚C/W and that using
Thermalloy Thermacote thermal compound provides a ther-
mal resistance, θ
CS
(case to heat sink), of about 0.2˚C/W as
explained in the Heat Sinking section.
Referring to the figure below, it is seen that the thermal resis-
tance from the die (junction) to the outside air (ambient) is a
combination of three thermal resistances, two of which are
known, θ
JC
and θ
CS
. Since convection heat flow (power dis-
sipation) is analogous to current flow, thermal resistance is
analogous to electrical resistance, and temperature drops
are analogous to voltage drops, the power dissipation out of
the LM3886 is equal to the following:
P
DMAX
=(T
Jmax
−T
Amb
)/θ
JA
where θ
JA
=θ
JC
+θ
CS
+θ
SA
But since we know P
DMAX
,θ
JC
, and θ
SC
for the application
and we are looking for θ
SA
, we have the following:
θ
SA
=[(T
Jmax
−T
Amb
)−P
DMAX
(θ
JC
+θ
CS
)]/P
DMAX
(4)
Again it must be noted that the value of θ
SA
is dependent
upon the system designer’s amplifier application and its cor-
responding parameters as described previously. If the ambi-
ent temperature that the audio amplifier is to be working un-
der is higher than the normal 25˚C, then the thermal
resistance for the heat sink, given all other things are equal,
will need to be smaller.
Equations (1), (4)
are the only equations needed in the de-
termination of the maximum heat sink thermal resistance.
This is of course given that the system designer knows the
required supply voltages to drive his rated load at a particular
power output level and the parameters provided by the semi-
conductor manufacturer. These parameters are the junction
to case thermal resistance, θ
JC
,T
Jmax
=150˚C, and the rec-
ommended Thermalloy Thermacote thermal compound re-
sistance, θ
CS
.
SIGNAL-TO-NOISE RATIO
In the measurement of the signal-to-noise ratio, misinterpre-
tations of the numbers actually measured are common. One
amplifier may sound much quieter than another, but due to
improper testing techniques, they appear equal in measure-
ments. This is often the case when comparing integrated cir-
cuit designs to discrete amplifier designs. Discrete transistor
amps often “run out of gain” at high frequencies and there-
fore have small bandwidths to noise as indicated below.
Integrated circuits have additional open loop gain allowing
additional feedback loop gain in order to lower harmonic dis-
tortion and improve frequency response. It is this additional
bandwidth that can lead to erroneous signal-to-noise mea-
surements if not considered during the measurement pro-
cess. In the typical example above, the difference in band-
width appears small on a log scale but the factor of 10 in
bandwidth, (200 kHz to 2 MHz) can result in a 10 dB theoreti-
cal difference in the signal-to-noise ratio (white noise is pro-
portional to the square root of the bandwidth in a system).
In comparing audio amplifiers it is necessary to measure the
magnitude of noise in the audible bandwidth by using a
“weighting” filter (Note 18). A “weighting” filter alters the fre-
quency response in order to compensate for the average hu-
man ear’s sensitivity to the frequency spectra. The weighting
filters at the same time provide the bandwidth limiting as dis-
cussed in the previous paragraph.
Note 18: CCIR/ARM:
A Practical Noise Measurement Method;
by Ray
Dolby, David Robinson and Kenneth Gundry, AES Preprint No. 1353 (F-3).
In addition to noise filtering, differing meter types give differ-
ent noise readings. Meter responses include:
1. RMS reading,
2. average responding,
3. peak reading, and
4. quasi peak reading.
Although theoretical noise analysis is derived using true
RMS based calculations, most actual measurements are
taken with ARM (Average Responding Meter) test equip-
ment.
Typical signal-to-noise figures are listed for anA-weighted fil-
ter which is commonly used in the measurement of noise.
The shape of all weighting filters is similar, with the peak of
the curve usually occurring in the 3 kHz–7 kHz region as
shown below.
DS011833-12
DS011833-13
DS011833-14
www.national.com 14
Application Information (Continued)
SUPPLY BYPASSING
The LM3886 has excellent power supply rejection and does
not require a regulated supply. However, to eliminate pos-
sible oscillations all op amps and power op amps should
have their supply leads bypassed with low-inductance ca-
pacitors having short leads and located close to the package
terminals. Inadequate power supply bypassing will manifest
itself by a low frequency oscillation known as “motorboating”
or by high frequency instabilities. These instabilities can be
eliminated through multiple bypassing utilizing a large tanta-
lum or electrolytic capacitor (10 µF or larger) which is used to
absorb low frequency variations and a small ceramic capaci-
tor (0.1 µF) to prevent any high frequency feedback through
the power supply lines.
If adequate bypassing is not provided the current in the sup-
ply leads which is a rectified component of the load current
may be fed back into internal circuitry. This signal causes low
distortion at high frequencies requiring that the supplies be
bypassed at the package terminals with an electrolytic ca-
pacitor of 470 µF or more.
LEAD INDUCTANCE
Power op amps are sensitive to inductance in the output
lead, particularly with heavy capacitive loading. Feedback to
the input should be taken directly from the output terminal,
minimizing common inductance with the load.
Lead inductance can also cause voltage surges on the sup-
plies. With long leads to the power supply, energy is stored in
the lead inductance when the output is shorted. This energy
can be dumped back into the supply bypass capacitors when
the short is removed. The magnitude of this transient is re-
duced by increasing the size of the bypass capacitor near
the IC. With at least a 20 µF local bypass, these voltage
surges are important only if the lead length exceeds a couple
feet (>1 µH lead inductance). Twisting together the supply
and ground leads minimizes the effect.
LAYOUT, GROUND LOOPS AND STABILITY
The LM3886 is designed to be stable when operated at a
closed-loop gain of 10 or greater, but as with any other
high-current amplifier, the LM3886 can be made to oscillate
under certain conditions. These usually involve printed cir-
cuit board layout or output/input coupling.
When designing a layout, it is important to return the load
ground, the output compensation ground, and the low level
(feedback and input) grounds to the circuit board common
ground point through separate paths. Otherwise, large cur-
rents flowing along a ground conductor will generate volt-
ages on the conductor which can effectively act as signals at
the input, resulting in high frequency oscillation or excessive
distortion. It is advisable to keep the output compensation
components and the 0.1 µF supply decoupling capacitors as
close as possible to the LM3886 to reduce the effects of PCB
trace resistance and inductance. For the same reason, the
ground return paths should be as short as possible.
In general, with fast, high-current circuitry, all sorts of prob-
lems can arise from improper grounding which again can be
avoided by returning all grounds separately to a common
point. Without isolating the ground signals and returning the
grounds to a common point, ground loops may occur.
“Ground Loop” is the term used to describe situations occur-
ring in ground systems where a difference in potential exists
between two ground points. Ideally a ground is a ground, but
unfortunately, in order for this to be true, ground conductors
with zero resistance are necessary. Since real world ground
leads possess finite resistance, currents running through
them will cause finite voltage drops to exist. If two ground re-
turn lines tie into the same path at different points there will
be a voltage drop between them. The first figure below
shows a common ground example where the positive input
ground and the load ground are returned to the supply
ground point via the same wire. The addition of the finite wire
resistance, R
2
, results in a voltage difference between the
two points as shown below.
The load current I
L
will be much larger than input bias current
I
I
, thus V
1
will follow the output voltage directly, i.e. in phase.
Therefore the voltage appearing at the non-inverting input is
effectively positive feedback and the circuit may oscillate. If
there were only one device to worry about then the values of
R
1
and R
2
would probably be small enough to be ignored;
however, several devices normally comprise a total system.
Any ground return of a separate device, whose output is in
phase, can feedback in a similar manner and cause instabili-
ties. Out of phase ground loops also are troublesome, caus-
ing unexpected gain and phase errors.
The solution to most ground loop problems is to always use
a single-point ground system, although this is sometimes im-
practical. The third figure below is an example of a
single-point ground system.
The single-point ground concept should be applied rigor-
ously to all components and all circuits when possible. Viola-
tions of single-point grounding are most common among
printed circuit board designs, since the circuit is surrounded
by large ground areas which invite the temptation to run a
device to the closest ground spot. As a final rule, make all
ground returns low resistance and low inductance by using
large wire and wide traces.
Occasionally, current in the output leads (which function as
antennas) can be coupled through the air to the amplifier in-
put, resulting in high-frequency oscillation. This normally
happens when the source impedance is high or the input
leads are long. The problem can be eliminated by placing a
DS011833-15
www.national.com15
Application Information (Continued)
small capacitor, C
C
, (on the order of 50 pF to 500 pF) across
the LM3886 input terminals. Refer to the External Compo-
nents Description section relating to component interaction
with C
f
.
REACTIVE LOADING
It is hard for most power amplifiers to drive highly capacitive
loads very effectively and normally results in oscillations or
ringing on the square wave response. If the output of the
LM3886 is connected directly to a capacitor with no series
resistance, the square wave response will exhibit ringing if
the capacitance is greater than about 0.2 µF. If highly capaci-
tive loads are expected due to long speaker cables, a
method commonly employed to protect amplifiers from low
impedances at high frequencies is to couple to the load
through a 10Ωresistor in parallel with a 0.7 µH inductor. The
inductor-resistor combination as shown in the Typical Appli-
cation Circuit isolates the feedback amplifier from the load
by providing high output impedance at high frequencies thus
allowing the 10Ωresistor to decouple the capacitive load and
reduce the Q of the series resonant circuit. The LR combina-
tion also provides low output impedance at low frequencies
thus shorting out the 10Ωresistor and allowing the amplifier
to drive the series RC load (large capacitive load due to long
speaker cables) directly.
GENERALIZED AUDIO POWER AMPLIFIER DESIGN
The system designer usually knows some of the following
parameters when starting an audio amplifier design:
Desired Power Output Input Level
Input Impedance Load Impedance
Maximum Supply Voltage Bandwidth
The power output and load impedance determine the power
supply requirements, however, depending upon the applica-
tion some system designers may be limited to certain maxi-
mum supply voltages. If the designer does have a power
supply limitation, he should choose a practical load imped-
ance which would allow the amplifier to provide the desired
output power, keeping in mind the current limiting capabili-
ties of the device. In any case, the output signal swing and
current are found from (where P
O
is the average output
power):
(5)
(6)
To determine the maximum supply voltage the following pa-
rameters must be considered. Add the dropout voltage (4V
for LM3886) to the peak output swing, V
opeak
, to get the sup-
ply rail value (i.e. ±(V
opeak
+ Vod) at a current of I
opeak
). The
regulation of the supply determines the unloaded voltage,
usually about 15%higher. Supply voltage will also rise 10%
during high line conditions. Therefore, the maximum supply
voltage is obtained from the following equation:
Max. supplies )±(V
opeak
+ Vod)(1 + regulation)(1.1)(7)
The input sensitivity and the output power specs determine
the minimum required gain as depicted below:
(8)
Normally the gain is set between 20 and 200; for a 40W, 8Ω
audio amplifier this results in a sensitivity of 894 mV and
89 mV, respectively. Although higher gain amplifiers provide
greater output power and dynamic headroom capabilities,
there are certain shortcomings that go along with the so
called “gain.” The input referred noise floor is increased and
hence the SNR is worse. With the increase in gain, there is
also a reduction of the power bandwidth which results in a
decrease in feedback thus not allowing the amplifier to re-
spond quickly enough to nonlinearities. This decreased abil-
ity to respond to nonlinearities increases the THD + N speci-
fication.
The desired input impedance is set by R
IN
. Very high values
can cause board layout problems and DC offsets at the out-
put. The value for the feedback resistance, R
f1
, should be
chosen to be a relatively large value (10 kΩ–100 kΩ), and
the other feedback resistance, Ri, is calculated using stan-
dard op amp configuration gain equations. Most audio ampli-
fiers are designed from the non-inverting amplifier configura-
tion.
DESIGN A 40W/4ΩAUDIO AMPLIFIER
Given:
Power Output 40W
Load Impedance 4Ω
Input Level 1V(max)
Input Impedance 100 kΩ
Bandwidth 20 Hz–20 kHz ±0.25 dB
Equations (5), (6)
give:
40W/4ΩV
opeak
=17.9V I
opeak
=4.5A
Therefore the supply required is: ±21.0V @4.5A
With 15%regulation and high line the final supply voltage is
±26.6V using
Equation (7)
. At this point it is a good idea to
check the Power Output vs Supply Voltage to ensure that the
required output power is obtainable from the device while
maintaining low THD + N. It is also good to check the Power
Dissipation vs Supply Voltage to ensure that the device can
handle the internal power dissipation. At the same time de-
signing in a relatively practical sized heat sink with a low
thermal resistance is also important. Refer to Typical Per-
formance Characteristics graphs and the Thermal Con-
siderations section for more information.
The minimum gain from
Equation (8)
is: A
V
≥12.6
We select a gain of 13 (Non-Inverting Amplifier); resulting in
a sensitivity of 973 mV.
Letting R
IN
equal 100 kΩgives the required input imped-
ance, however, this would eliminate the “volume control” un-
less an additional input impedance was placed in series with
the 10 kΩpotentiometer that is depicted in
Figure 1
. Adding
the additional 100 kΩresistor would ensure the minumum
required input impedance.
For low DC offsets at the output we let R
f1
=100 kΩ. Solving
for Ri (Non-Inverting Amplifier) gives the following:
Ri =R
f1
/(A
V
−1)=100k/(13 − 1) =8.3 kΩ; use 8.2 kΩ
The bandwidth requirement must be stated as a pole, i.e.,
the 3 dB frequency. Five times away from a pole gives
0.17 dB down, which is better than the required 0.25 dB.
Therefore: f
L
=20 Hz/5 =4Hz
f
H
=20 kHz x 5 =100 kHz
At this point, it is a good idea to ensure that the
Gain-Bandwidth Product for the part will provide the de-
signed gain out to the upper 3 dB point of 100 kHz. This is
why the minimum GBWP of the LM3886 is important.
www.national.com 16
Application Information (Continued)
GBWP ≥A
V
xf3dB=13 x 100 kHz =1.3 MHz
GBWP =2.0 MHz (min) for the LM3886
Solving for the low frequency roll-off capacitor, Ci, we have:
Ci ≥1/(2πRi f
L
)=4.85 µF; use 4.7 µF.
Definition of Terms
Input Offset Voltage: The absolute value of the voltage
which must be applied between the input terminals through
two equal resistances to obtain zero output voltage and cur-
rent.
Input Bias Current: The absolute value of the average of
the two input currents with the output voltage and current at
zero.
Input Offset Current: The absolute value of the difference
in the two input currents with the output voltage and current
at zero.
Input Common-Mode Voltage Range (or Input Voltage
Range): The range of voltages on the input terminals for
which the amplifier is operational. Note that the specifica-
tions are not guaranteed over the full common-mode voltage
range unless specifically stated.
Common-Mode Rejection: The ratio of the input
common-mode voltage range to the peak-to-peak change in
input offset voltage over this range.
Power Supply Rejection: The ratio of the change in input
offset voltage to the change in power supply voltages pro-
ducing it.
Quiescent Supply Current: The current required from the
power supply to operate the amplifier with no load and the
output voltage and current at zero.
Slew Rate: The internally limited rate of change in output
voltage with a large amplitude step function applied to the in-
put.
Class B Amplifier: The most common type of audio power
amplifier that consists of two output devices each of which
conducts for 180˚ of the input cycle. The LM3886 is a
Quasi-AB type amplifier.
Crossover Distortion: Distortion caused in the output stage
of a class B amplifier. It can result from inadequate bias cur-
rent providing a dead zone where the output does not re-
spond to the input as the input cycle goes through its zero
crossing point. Also for ICs an inadequate frequency re-
sponse of the output PNP device can cause a turn-on delay
giving crossover distortion on the negative going transition
through zero crossing at the higher audio frequencies.
THD+N:Total Harmonic Distortion plus Noise refers to the
measurement technique in which the fundamental compo-
nent is removed by a bandreject (notch) filter and all remain-
ing energy is measured including harmonics and noise.
Signal-to-Noise Ratio: The ratio of a system’s output signal
level to the system’s output noise level obtained in the ab-
sence of a signal. The output reference signal is either speci-
fied or measured at a specified distortion level.
Continuous Average Output Power: The minimum sine
wave continuous average power output in watts (or dBW)
that can be delivered into the rated load, over the rated
bandwidth, at the rated maximum total harmonic distortion.
Music Power: A measurement of the peak output power ca-
pability of an amplifier with either a signal duration suffi-
ciently short that the amplifier power supply does not sag
during the measurement, or when high quality external
power supplies are used. This measurement (an IHF stan-
dard) assumes that with normal music program material the
amplifier power supplies will sag insignificantly.
Peak Power: Most commonly referred to as the power out-
put capability of an amplifier that can be delivered to the
load; specified by the part’s maximum voltage swing.
Headroom: The margin between an actual signal operating
level (usually the power rating of the amplifier with particular
supply voltages, a rated load value, and a rated THD + N fig-
ure) and the level just before clipping distortion occurs, ex-
pressed in decibels.
Large Signal Voltage Gain: The ratio of the output voltage
swing to the differential input voltage required to drive the
output from zero to either swing limit. The output swing limit
is the supply voltage less a specified quasi-saturation volt-
age.A pulse of short enough duration to minimize thermal ef-
fects is used as a measurement signal.
Output-Current Limit: The output current with a fixed out-
put voltage and a large input overdrive. The limiting current
drops with time once SPiKe protection circuitry is activated.
Output Saturation Threshold (Clipping Point): The output
swing limit for a specified input drive beyond that required for
zero output. It is measured with respect to the supply to
which the output is swinging.
Output Resistance: The ratio of the change in output volt-
age to the change in output current with the output around
zero.
Power Dissipation Rating: The power that can be dissi-
pated for a specified time interval without activating the pro-
tection circuitry. For time intervals in excess of 100 ms, dis-
sipation capability is determined by heat sinking of the IC
package rather than by the IC itself.
Thermal Resistance: The peak, junction-temperature rise,
per unit of internal power dissipation (units in ˚C/W), above
the case temperature as measured at the center of the pack-
age bottom.
The DC thermal resistance applies when one output transis-
tor is operating continuously. The AC thermal resistance ap-
plies with the output transistors conducting alternately at a
high enough frequency that the peak capability of neither
transistor is exceeded.
Power Bandwidth: The power bandwidth of an audio ampli-
fier is the frequency range over which the amplifier voltage
gain does not fall below 0.707 of the flat band voltage gain
specified for a given load and output power.
Power bandwidth also can be measured by the frequencies
at which a specified level of distortion is obtained while the
amplifier delivers a power output 3 dB below the rated out-
put. For example, an amplifier rated at 60W with ≤0.25%
THD + N, would make its power bandwidth measured as the
difference between the upper and lower frequencies at which
0.25%distortion was obtained while the amplifier was deliv-
ering 30W.
Gain-Bandwidth Product: The Gain-Bandwidth Product is
a way of predicting the high-frequency usefulness of an op
amp. The Gain-Bandwidth Product is sometimes called the
unity-gain frequency or unity-gain cross frequency because
the open-loop gain characteristic passes through or crosses
unity gain at this frequency. Simply, we have the following re-
lationship: A
CL1
xf
1
=A
CL2
xf
2
Assuming that at unity-gain (A
CL1
=1 or (0 dB)) fu =fi =
GBWP, then we have the following: GBWP =A
CL2
xf2
This says that once fu (GBWP) is known for an amplifier,
then the open-loop gain can be found at any frequency. This
www.national.com17
Definition of Terms (Continued)
is also an excellent equation to determine the 3 dB point of a
closed-loop gain, assuming that you know the GBWP of the
device. Refer to the diagram on the following page.
Biamplification: The technique of splitting the audio fre-
quency spectrum into two sections and using individual
power amplifiers to drive a separate woofer and tweeter.
Crossover frequencies for the amplifiers usually vary be-
tween 500 Hz and 1600 Hz. “Biamping” has the advantages
of allowing smaller power amps to produce a given sound
pressure level and reducing distortion effects produced by
overdrive in one part of the frequency spectrum affecting the
other part.
C.C.I.R./A.R.M.:
Literally: International Radio Consultative Committee
Average Responding Meter
This refers to a weighted noise measurement for a Dolby B
type noise reduction system. A filter characteristic is used
that gives a closer correlation of the measurement with the
subjective annoyance of noise to the ear. Measurements
made with this filter cannot necessarily be related to un-
weighted noise measurements by some fixed conversion
factor since the answers obtained will depend on the spec-
trum of the noise source.
S.P.L.: Sound Pressure Level—usually measured with a
microphone/meter combination calibrated to a pressure level
of 0.0002 µBars (approximately the threshold hearing level).
S.P.L. =20 Log 10P/0.0002 dB
where P is the R.M.S. sound pressure in microbars.
(1 Bar =1 atmosphere =14.5 lb/in
2
=194 dB S.P.L.).
DS011833-16
www.national.com 18
Physical Dimensions inches (millimeters) unless otherwise noted
Order Number LM3886T
NS Package Number TA11B
Order Number LM3886TF
NS Package Number TF11B
www.national.com19
Notes
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DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT AND GENERAL
COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein:
1. Life support devices or systems are devices or
systems which, (a) are intended for surgical implant
into the body, or (b) support or sustain life, and
whose failure to perform when properly used in
accordance with instructions for use provided in the
labeling, can be reasonably expected to result in a
significant injury to the user.
2. A critical component is any component of a life
support device or system whose failure to perform
can be reasonably expected to cause the failure of
the life support device or system, or to affect its
safety or effectiveness.
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www.national.com
LM3886
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