LT1769CGN#PBF - LINEAR TECHNOLOGY

LT1769

1769fa

Constant-Current/

Constant-Voltage 2A Battery

Charger with Input Current Limiting

■

Simple Solution to Charge NiCd, NiMH and Lithium

Rechargeable Batteries—Charging Current

Programmed by Resistors or DAC

■

Adapter Current Limit Allows Maximum Possible

Charging Current During System Use*

■

Precision 0.5% Accuracy for Voltage Mode Charging

■

Available in 20-Lead Exposed Pad TSSOP and

28-Lead Narrow SSOP Packages

■

High Efficiency Current Mode PWM with 3A Internal

Switch

■

5% Charge Current Accuracy

■

Adjustable Undervoltage Lockout

■

Automatic Shutdown When AC Adapter is Removed

■

Low Reverse Battery Drain Current: 3µA

■

Current Sensing Can Be at Either Terminal of the Battery

■

Charging Current Soft Start

■

Shutdown Control

The LT

1769 current mode PWM battery charger is a

simple, efficient solution to fast charge modern recharge-

able batteries including lithium-ion (Li-Ion), nickel-metal-

hydride (NiMH) and nickel-cadmium (NiCd) that require

constant-current and/or constant-voltage charging. The

internal switch is capable of delivering 2A** DC current

(3A peak current). Charge current can be programmed by

resistors or a DAC to within 5%. With 0.5% reference voltage

accuracy, the LT1769 meets the critical constant-voltage

charging requirement for Li-Ion cells.

A third control loop is provided to regulate the current

drawn from the input AC adapter. This allows simulta-

neous operation of the equipment and battery charging

without overloading the adapter. Charge current is reduced

to keep the adapter current below specified levels.

The LT1769 can charge batteries ranging from 1V to 20V.

Ground sensing of current is not required and the battery’s

negative terminal can be tied directly to ground. A saturat-

ing switch running at 200kHz gives high charging effi-

ciency and small inductor size. A blocking diode is not

required between the chip and the battery because the

chip goes into sleep mode and drains only 3µA when the

wall adapter is unplugged.

Figure 1. 2A Lithium-Ion Battery Charger

■

Chargers for NiCd, NiMH, Lead-Acid, Lithium

Rechargeable Batteries

■

Switching Regulators with Precision Current Limit

, LTC and LT are registered trademarks of Linear Technology Corporation.

*US patent number 5,723,970

**See LT1510 for 1.5A charger; see LT1511 for 3A charger

FEATURES

DESCRIPTIO

APPLICATIO S

TYPICAL APPLICATIO

BOOST

COMP1

CLN

OVP SENSE BAT

1µF

RS4†

ADAPTER

CURRENT SENSE

R7†

500Ω

R5†

UNDERVOLTAGE

LOCKOUT

VIN (ADAPTER INPUT)

11V TO 28V

VBAT

CPROG

1µF

CIN*

15µF

300ΩRPROG

4.93k

0.33µF

0.47µF

RS3

200Ω

RS2

200Ω

L1**

22µHD2

1N4148 2nF

10k

RS1

0.05Ω

BATTERY CURRENT

SENSE

390k

0.25%

BATTERY

VOLTAGE SENSE

162k

0.25%

COUT

22µF

TANT

8.4V

Li-Ion

LT1769

NOTE: COMPLETE LITHIUM-ION CHARGER,

NO TERMINATION REQUIRED. RS4, R7

AND C1 ARE OPTIONAL FOR IIN LIMITING

*TOKIN OR UNITED CHEMI-CON/MARCON

CERAMIC SURFACE MOUNT

**22µH SUMIDA CDRH125

†SEE APPLICATIONS INFORMATION FOR

INPUT CURRENT LIMIT AND UNDERVOLTAGE LOCKOUT

††GENERAL SEMICONDUCTOR. FOR TJ LESS THEN 100°C

MBRS130LT3 CAN BE USED

VCC TO MAIN

SYSTEM LOAD

SPIN

D1††

SS24

GND CLP

D3††

SS24

1511 • F01

PROG

LT1769

1769fa

ABSOLUTE MAXIMUM RATINGS

PACKAGE/ORDER INFORMATION

ORDER PART

NUMBER

*ALL VCC PINS SHOULD

BE CONNECTED

TOGETHER CLOSE TO

THE PINS

** ALL GND PINS ARE

FUSED TO INTERNAL DIE

ATTACH PADDLE FOR

HEAT SINKING. CONNECT

THESE PINS TO

EXPANDED PC LANDS

FOR PROPER HEAT

SINKING. 35°C/W

THERMAL RESISTANCE

ASSUMES AN INTERNAL

GROUND PLANE

DOUBLING AS A HEAT

SPREADER

LT1769CGN

LT1769IGN

JMAX

= 125°C, θ

= 35°C/ W**

TOP VIEW

GN PACKAGE

28-LEAD PLASTIC SSOP

GND**

BOOST

GND**

OVP

CLP

CLN

COMP1

SENSE

GND**

CC1

CC2

CC3

GND**

PROG

OUT

COMP2

BAT

SPIN

GND**

Supply Voltage

, CLP and CLN Pin Voltage)......................... 30V

BOOST Pin Voltage with Respect to V

................. 25V

BAT

(Average)........................................................... 2A

Operating Junction Temperature Range

Commercial ........................................... 0°C to 125°C

Industrial ......................................... –40°C to 125°C

Consult LTC Marketing for parts specified with wider operating temperature ranges.

(Note 1)

Operating Ambient Temperature

Commercial ............................................ 0°C to 70°C

Industrial ........................................... –40°C to 85°C

Storage Temperature Range ................. –65°C to 150°C

Lead Temperature (Soldering, 10 sec)..................300°C

ORDER PART

NUMBER

LT1769CFE

LT1769IFE

FE PACKAGE

20-LEAD PLASTIC TSSOP

TOP VIEW

BOOST

GND

OVP

CLN

CLP

COMP1

SENSE

GND

CC1

CC2

CC3

PROG

GND

OUT

BAT

SPIN

JMAX

= 125°C, θ

= 35°C/W

†

THE BOTTOM METAL

PLATE OF THIS PACKAGE

IS FUSED TO INTERNAL

GROUND AND IS FOR

HEAT SINKING. SOLDER

THE BOTTOM METAL

PLATE ONTO PCB

GROUND PLANE FOR

HEAT SINKING.

ELECTRICAL CHARACTERISTICS

The ● denotes specifications which apply over the full operating

temperature range, otherwise specifications are at TA = 25°C. VCC = 16V, VBAT = 8V, RS2 = RS3 = 200Ω (see Block Diagram),

VCLN = VCC. No load on any outputs unless otherwise noted.

PARAMETER CONDITIONS MIN TYP MAX UNITS

Overall

Supply Current V

PROG

= 2.7V, V

≤ 20V ●4.5 6.8 mA

PROG

= 2.7V, 20V < V

≤ 25V ●4.6 7.0 mA

Sense Amplifier CA1 Gain and Input Offset Voltage 8V ≤ V

≤ 25V , 0V ≤ V

BAT

≤ 20V

(With R

= 200Ω, R

= 200Ω)R

PROG

= 4.93k ●93 100 107 mV

(Measured across R

)(Note 2) R

PROG

= 49.3k ●81012 mV

< 0°C713mV

= 28V, V

BAT

= 20V

PROG

= 4.93k ●90 110 mV

PROG

= 49.3k ●713mV

< 0°C614mV

EXPOSED PAD IS GROUND

(MUST BE SOLDERED TO PCB)

EXPOSED PAD SIZE: 3.0

(.188) ×4.1

(.162)

LT1769

1769fa

ELECTRICAL CHARACTERISTICS

The ● denotes specifications which apply over the full operating

temperature range, otherwise specifications are at TA = 25°C. VCC = 16V, VBAT = 8V, RS2 = RS3 = 200Ω (see Block Diagram),

VCLN = VCC. No load on any outputs unless otherwise noted.

PARAMETER CONDITIONS MIN TYP MAX UNITS

Overall

Undervoltage Lockout (Switch OFF) Threshold Measured at UV Pin ●678 V

UV Pin Input Current 0.2V ≤ V

≤ 8V ●0.1 5 µA

UV Output Voltage at UV

OUT

Pin In Undervoltage State, I

UVOUT

= 70µA●0.1 0.5 V

UV Output Leakage Current at UV

OUT

Pin 8V ≤ V

, V

UVOUT

= 5V ●0.1 3 µA

Reverse Current from Battery (When V

Is V

BAT

≤ 20V, V

≤ 0.4V 3 15 µA

Not Connected, V

Is Floating)

Boost Pin Current V

= 20V, V

BOOST

= 0V 0.1 10 µA

= 28V, V

BOOST

= 0V 0.25 20 µA

2V ≤ V

BOOST

– V

< 8V (Switch ON) 6 9 mA

8V ≤ V

BOOST

– V

≤ 25V (Switch ON) 8 12 mA

Switch

Switch ON Resistance 8V ≤ V

≤ V

MAX

, I

= 2A,

BOOST

– V

≥ 2V ●0.15 0.25 Ω

∆I

BOOST

/∆I

During Switch ON V

BOOST

= 24V, I

≤ 2A 25 35 mA/A

Switch OFF Leakage Current V

= 0V, V

≤ 20V ●2 100 µA

20V < V

≤ 28V ●4 200 µA

Minimum I

PROG

for Switch ON 2420 µA

Minimum I

PROG

for Switch OFF ●1 2.4 mA

Maximum V

BAT

for Switch ON ●V

– 2 V

Current Sense Amplifier CA1 Inputs (Sense, BAT)

Input Bias Current ●–50 –125 µA

Input Common Mode Low ●–0.25 V

Input Common Mode High ●V

– 2 V

SPIN Input Current –100 –200 µA

Reference

Reference Voltage (Note 3) R

PROG

= 4.93k, Measured at OVP with

VA Supplying I

PROG

and Switch OFF 2.448 2.465 2.477 V

Reference Voltage All Conditions of V

≥ 0°C●2.441 2.489 V

< 0°C (Note 4) ●2.43 2.489 V

Oscillator

Switching Frequency 180 200 220 kHz

Switching Frequency All Conditions of V

≥ 0°C●170 200 230 kHz

< 0°C●160 230 kHz

Maximum Duty Cycle 90 93 %

●85 %

Current Amplifier CA2

Transconductance V

= 1V, I

= ±1µA 150 250 550 µmho

Maximum V

for Switch OFF ●0.6 V

Current (Out of Pin) V

≥ 0.6V 100 µA

< 0.45V 3 mA

LT1769

1769fa

ELECTRICAL CHARACTERISTICS

TYPICAL PERFORMANCE CHARACTERISTICS

Efficiency of Figure 1 Circuit

BAT

(A)

0.2

EFFICIENCY (%)

100

80 1.0 1.8 2.2

1769 G01

0.6 1.4

= 16.5

BAT

= 8.4V

CHARGER EFFICIENCY

INCLUDES LOSS

IN DIODE D3

VCC (V)

ICC (mA)

7.0

6.5

6.0

5.5

5.0

4.5 510 15 20

1769 G03

25 30

TJ = 125°C

TJ = 25°C

TJ = 0°C

MAXIMUM DUTY CYCLE

ICC vs Duty Cycle

DUTY CYCLE (%)

010305070

(mA)

1769 G02

20 40 60

= 125°C

= 0°C

= 25°C

= 16V

ICC vs V

The ● denotes specifications which apply over the full operating

temperature range, otherwise specifications are at TA = 25°C. VCC = 16V, VBAT = 8V. No load on any outputs unless otherwise noted.

Note 1: Absolute Maximum Ratings are those values beyond which the life

of a device may be impaired.

Note 2: Tested with Test Circuit 1.

Note 3: Tested with Test Circuit 2.

Note 4: A linear interpolation can be used for reference voltage

specification between 0°C and –40°C.

PARAMETER CONDITIONS MIN TYP MAX UNITS

Voltage Amplifier VA

Transconductance (Note 3) Output Current from 50µA to 500µA 0.25 0.6 1.3 mho

Output Source Current V

OVP

= V

REF

+ 10mV, V

PROG

= V

REF

+ 10mV 1.1 mA

OVP Input Bias Current VA Output Current at 0.5mA ●±3±10 nA

VA Output Current at 0.5mA, T

> 90°C●–15 25 nA

VA Output Current at 0.5mA, T

< 0°C±15 nA

Current Limit Amplifier CL1, 8V

≤

Input Common Mode

Turn-On Threshold 0.5mA Output Current 93 100 110 mV

Transconductance Output Current from 50µA to 500µA 0.5 1 2 mho

CLP Input Current 0.5mA Output Current, V

≥ 0.4V 0.3 1 µA

CLN Input Current 0.5mA Output Current V

≥ 0.4V 0.8 2 mA

LT1769

1769fa

TYPICAL PERFORMANCE CHARACTERISTICS

(V)

∆V

REF

(V)

0.003

0.002

0.001

–0.001

–0.002

–0.003 510 15 20

1769 G04

25 30

ALL TEMPERATURES

VREF Line Regulation IVA vs ∆VOVP (Voltage Amplifier)

(mA)

∆V

OVP

(mV)

00.8

1769 G05

0.20.1 0.3 0.5 0.7 0.9

0.4 0.6 1.0

= 125°C

= 25°C

VC Pin Characteristics

(V)

0 0.2 0.6 1.0 1.4 1.8

(mA)

–1.20

–1.08

–0.96

–0.84

–0.72

–0.60

–0.48

–0.36

–0.24

–0.12

0.12 1.6

1769 G07

0.4 0.8 1.2 2.0

JUNCTION TEMPERATURE (°C)

DUTY CYCLE (%)

120

1769 G06

40 80

90 20 60 100 140

Maximum Duty Cycle

Reference Voltage

vs Temperature

JUNCTION TEMPERATURE

REFERENCE VOLTAGE (V)

2.470

2.468

2.466

2.464

2.462

2.460

2.458 25 50 75 100

1769 G09

125 150

PROG

(V)

0123 54

PROG

(mA)

–6

1769 G08

= 125°C

= 25°C

PROG Pin Characteristics

LT1769

1769fa

GND (Pins 1 to 3, 7, 8, 14, 15, 22, 26 to 28/Pins 4, 5, 14,

20): Ground Pins. Must be connected to expanded PC

lands for proper heat sinking. See Applications Informa-

tion section for details.

SW (Pin 4/Pin 1): Switch Output. The Schottky catch

diode must be placed with very short lead length in close

proximity to SW pin and GND.

BOOST (Pin 5/Pin 2): This pin is used to bootstrap and

drive the switch power NPN transistor to a low on-voltage

for low power dissipation. In Figure 1, V

BOOST

= V

+ V

BAT

when switch is on. For lowest IC power dissipation,

connect boost diode D1 to a 3V to 6V at 30mA voltage

source (see Figure 10).

UV (Pin 6/Pin 3): Undervoltage Lockout Input. The rising

threshold is at 6.7V with a hysteresis of 0.5V. Switching

stops in undervoltage lockout. When the input supply

(normally the wall adapter output) to the IC is removed, the

UV pin must be pulled down to below 0.7V (a 5k resistor

from adapter output to GND is required) otherwise the

reverse battery current drained by the IC will be approxi-

mately 200µA instead of 3µA. Do not leave the UV pin

floating. When connected to V

with no resistor divider,

the built-in 6.7V undervoltage lockout will be effective.

OVP (Pin 9/Pin 6): This is the input to amplifier VA with a

threshold of 2.465V. Typical bias current is about 3nA out

of this pin. For charging lithium-ion batteries, VA monitors

the battery voltage and reduces charging when battery

voltage reaches the preset value. If it is not used, the OVP

pin should be grounded.

CLP (Pin 10/Pin 8): This is the positive input to the input

current limit amplifier CL1. The threshold is set at 100mV.

When used to limit supply current, a filter is needed to filter

out the 200kHz switching noise.

CLN (Pin 11/Pin 7): This is the negative input to the input

current limit amplifier CL1.

COMP1 (Pin 12/Pin 9): This is the compensation node for

the input current limit amplifier CL1. At input adapter

current limit, this node rises to 1V. By forcing COMP1 low

with an external transistor, amplifier CL1 will be defeated

(no adapter current limit). COMP1 can source 200µA. If

this function is not used, the resistor and capacitor on

COMP1 pin, shown on the Figure 1 circuit, are not needed.

SENSE (Pin 13/Pin 10): Current Amplifier CA1 Input.

Sensing can be at either terminal of the battery.

SPIN (Pin 16/Pin 11): This pin is for the current amplifier

CA1 bias. It must be connected to R

as shown in the 2A

Lithium Battery Charger (Figure 1).

BAT (Pin 17/Pin 12): Current Amplifier CA1 Input.

COMP2 (Pin 18): This is also a compensation node for

amplifier CL1. Voltage on this pin rises to 2.8V at input

adapter current limit and/or at constant-voltage charging.

OUT

(Pin 19/Pin 13): This is an open-collector output for

undervoltage lockout status. It stays low in undervoltage

state. With an external pull-up resistor, it goes high at valid

. Note that the base drive of the open-collector NPN

comes from CLN pin. UV

OUT

stays low only when CLN is

higher than 2V. Pull-up current should be kept under 100µA.

(Pin 20/Pin 15): This is the inner loop control signal for

the current mode PWM. Switching starts at 0.7V. In

normal operation, a higher V

corresponds to higher

charge current. A capacitor of at least 0.33µF to GND

filters out noise and controls the rate of soft start. To stop

switching, pull this pin low. Typical output current is 30µA.

PROG (Pin 21/Pin 16): This pin is for programming the

charge current and for system loop compensation. During

normal operation, V

PROG

stays close to 2.465V. If it is

shorted to GND switching will stop. When a microproces-

sor controlled DAC is used to program charge current, it

must be capable of sinking current at a compliance up to

2.465V.

CC1

CC2

CC3

(Pins 23 to 25/Pins 17 to 19): Input

Supply. For good bypass, a low ESR capacitor of 15µF or

higher is required, with the lead length kept to a minimum.

should be between 8V and 28V and at least 3V higher

than V

BAT

. Undervoltage lockout starts and switching

stops when V

goes below 7V typical. Note that there is

an internal parasitic diode from SW pin to V

pin. Do not

force V

below SW by more than 0.7V with battery

present. All three V

pins should be shorted together

close to the pins.

PIN FUNCTIONS

UUU

(GN/FE Pin Numbers)

LT1769

1769fa

BLOCK DIAGRAM

–

VSW

0.7V

1.5V

VBAT

VREF

GND

SLOPE COMPENSATION

PWM

CA2

–

CA1

–

VREF

2.465V

SHUTDOWN

200kHz

OSCILLATOR

RPROG

VCC

UVOUT

VCC

BOOST

SENSE

SPIN

BAT

IPROG

RS3

RS2 RS1

IBAT

0VP

BAT

1769 BD

PROG

IPROG

IBAT = (IPROG)(RS2)

RS1

CPROG

75k

QSW

VCC

gm = 0.64

Ω

–

CL1

CLP

100mV

CLN

COMP1

COMP2

(RS3 = RS2)

2.465V

RPROG

RS2

RS1

(())

LT1769

1769fa

TEST CIRCUITS

Test Circuit 1

–

VREF

≈ 0.65V

VBAT

VCCA2

–

CA1

300Ω

20k

RS1

100Ω

BAT

SENSE

SPIN

1769 TC01

PROG

RPROG

0.047µF

LT1769

1µF

60k

LT1006

RS2

200Ω

RS3

200Ω

Test Circuit 2

REF

2.465V

–

10k

OVP

1769 TC02

PROG

LT1769

PROG

LT1013

0.47µF

–

OPERATION

The LT1769 is a current mode PWM step-down (buck)

switcher. The battery DC charge current is programmed

by a resistor R

PROG

(or a DAC output current) at the PROG

pin (see Block Diagram). Amplifier CA1 converts the

charge current through R

to a much lower current I

PROG

fed into the PROG pin. Amplifier CA2 compares the output

of CA1 with the programmed current and drives the PWM

control loop to force them to be equal. High DC accuracy

is achieved with averaging capacitor C

PROG

. Note that

PROG

has both AC and DC components. I

PROG

goes

through R1 and generates a ramp signal that is fed to the

PWM control comparator C1 through buffer B1 and level

shift resistors R2 and R3, forming the current mode inner

loop. The BOOST pin drives the switch NPN Q

into

saturation and reduces power loss. For batteries like

lithium-ion that require both constant-current and con-

stant-voltage charging, the 0.5%, 2.465V reference and

the amplifier VA reduce the charge current when battery

voltage reaches the preset level. For NiMH and NiCd, VA

can be used for overvoltage protection. When the input

voltage is removed, the V

pin drops to 0.7V below the

battery voltage, forcing the charger into a low battery drain

(3µA typical) sleep mode. To shut down the charger,

simply pull the V

pin low with a transistor.

LT1769

1769fa

APPLICATIONS INFORMATION

WUUU

Input and Output Capacitors

In the 2A Lithium-Ion Battery Charger (Figure 1), the input

capacitor (C

) is assumed to absorb all input switching

ripple current in the converter, so it must have adequate

ripple current rating. Worst-case RMS ripple current will

be equal to one half of the output charge current. Actual

capacitance value is not critical. Solid tantalum capacitors

such as the AVX TPS and Sprague 593D series have high

ripple current rating in a relatively small surface mount

package, but

caution must be used when tantalum capaci-

tors are used for input bypass

. High input surge currents

are possible when the adapter is hot-plugged to the

charger and solid tantalum capacitors have a known

failure mechanism when subjected to very high turn-on

surge currents. Selecting a high voltage rating on the

capacitor will minimize problems. Consult with the manu-

facturer before use. Alternatives include new high capacity

ceramic (5µF to 20µF) from Tokin or United Chemi-Con/

Marcon, et al. Sanyo OS-CON can also be used.

The output capacitor (C

OUT

) is also assumed to absorb

output switching ripple current. The general formula for

capacitor ripple current is:

RMS

(L1)(f)

BAT

()

0.29 (V

BAT

) 1 –

For example, V

= 16V, V

BAT

= 8.4V, L1 = 20µH,

and f = 200kHz, I

RMS

= 0.3A.

EMI considerations usually make it desirable to minimize

ripple current in the battery leads. Beads or inductors can

be added to increase battery impedance at the 200kHz

switching frequency. Switching ripple current splits be-

tween the battery and the output capacitor depending on

the ESR of the output capacitor and the battery imped-

ance. If the ESR of C

OUT

is 0.2Ω and the battery impedance

is raised to 4Ω with a bead or inductor, only 5% of the

ripple current will flow into the battery.

Soft-Start and Undervoltage Lockout

The LT1769 is soft-started by the 0.33µF capacitor on the

pin. On start-up, the V

pin voltage will quickly rise to

0.5V, then ramp at a rate set by the internal 45µA pull-up

current and the external capacitor. Charge current starts

ramping up when V

pin voltage reaches 0.7V and full

current is achieved with V

at 1.1V. With a 0.33µF capaci-

tor, the time to reach full charge current is about 10ms and

it is assumed that input voltage to the charger will reach full

value in less than 10ms. The capacitor can be

increased up to 1µF if longer input start-up times are needed.

In any switching regulator, conventional time-based soft-

starting can be defeated if the input voltage rises much

slower than the time out period. This happens because the

switching regulators in the battery charger and the com-

puter power supply are typically supplying a fixed amount

of power to the load. If the input voltage comes up slowly

compared to the soft-start time, the regulators will try to

deliver full power to the load when the input voltage is still

well below its final value. If the adapter is current limited,

it cannot deliver full power at reduced output voltages and

the possibility exists for a quasi “latch” state where the

adapter output stays in a current limited state at reduced

output voltage. For instance, if maximum charger plus

computer load power is 25W, a 15V adapter might be

current limited at 2A. If adapter voltage is less than

(25W/2A = 12.5V) when full power is drawn, the adapter

voltage will be pulled down by the constant 25W load until

it reaches a lower stable state where the switching regu-

lators can no longer supply full load. This situation can be

prevented by utilizing

undervoltage lockout

, set higher than

the minimum adapter voltage where full power can be

achieved.

A fixed undervoltage lockout of 7V is built into the LT1769.

This 7V threshold can be increased by adding a resistive

divider to the UV pin as shown in Figure 2. Internal lockout

is performed by clamping the V

pin low. The V

pin is

released from its clamped state when the UV pin rises

above 7V and is pulled low when the UV pin drops below

6.5V (0.5V hysteresis). At the same time UV

OUT

goes high

with an external pull-up resistor. This signal can be used

to alert the system that charging is about to start. The

charger will start delivering current about 4ms after V

released, as set by the 0.33µF capacitor. A resistor divider

is used to set the desired V

lockout voltage as shown in

Figure 2. A typical value for R6 is 5k and R5 is found from:

R5 = R6(V – V )

LT1769

1769fa

APPLICATIONS INFORMATION

WUUU

= Rising lockout threshold on the UV pin

= Charger input voltage that will sustain full load power

Example: With R6 = 5k, V

= 6.7V and setting V

at 12V;

R5 = 5k (12V – 6.7V)/6.7V = 4k

The resistor divider should be connected directly to the

adapter output as shown, not to the V

pin, to prevent

battery drain with no adapter voltage. If the UV pin is not

used, connect it to the adapter output (not V

) and

connect a resistor no greater than 5k to ground. Floating

this pin will cause reverse battery current to increase from

3µA to 200µA.

If connecting the unused UV pin to the adapter output is not

possible, it can be grounded. Although it would seem that

grounding the pin creates a permanent lockout state, the

UV circuitry is arranged for phase reversal with low volt-

ages on the UV pin to allow the grounding technique to work.

adapter load current remains below the limit. Amplifier

CL1 in Figure 2 senses the voltage across R

, connected

between the CLP and CLN pins. When this voltage exceeds

100mV, the amplifier will override the programmed charge

current to limit adapter current to 100mV/R

. A lowpass

filter formed by 500Ω and 1µF is required to eliminate

switching noise. If the input current limit is not used, both

CLP and CLN pins should be connected to V

Charge Current Programming

The basic formula for charge current is (see Block

Diagram):

IBAT = IPROG = 2.465V

RPROG

RS2

RS1

()()

RS2

RS1

()

where R

PROG

is the total resistance from PROG pin to ground.

For the sense amplifier CA1 biasing purpose, R

should

have the same value as R

and SPIN should be connected

directly to the sense resistor (R

) as shown in the Block

Diagram.

For example, 2A charge current is needed. For low power

dissipation on R

and enough signal to drive the amplifier

CA1, let R

= 100mV/2A = 0.05Ω. This limits R

power

to 0.2W. Let R

PROG

= 5k, then:

= R

= = 200Ω

BAT

)(R

PROG

)(R

)

2.465V

(2A)(5k)(0.05)

2.465V

Charge current can also be programmed by pulse width

modulating I

PROG

with a switch Q1 to R

PROG

at a frequency

higher than a few kHz (Figure 3). Charge current will be

proportional to the duty cycle of the switch with full current

at 100% duty cycle.

Lithium-Ion Charging

The 2A Lithium-Ion Battery Charger (Figure 1) charges at

a constant 2A until battery voltage reaches a limit set by R3

and R4. The charger will then automatically go into a

constant-voltage mode with current decreasing to near

zero over time as the battery reaches full charge. This is the

normal regimen for lithium-ion charging, with the charger

100mV

500Ω

CLP

CLN

1769 F02

LT1769

1µF

AC ADAPTER

OUTPUT

= 100mV

ADAPTER CURRENT LIMIT

–

CL1

Figure 2. Adapter Input Current Limiting

Adapter Current Limiting

An important feature of the LT1769 is the ability to

automatically adjust charge current to a level which avoids

overloading the wall adapter. This allows the product to

operate at the same time the batteries are being charged

without complex load management algorithms. Addition-

ally, batteries will automatically be charged at the maximum

possible rate of which the adapter is capable.

This is accomplished by sensing total adapter output

current and adjusting the charge current downward if a

preset adapter current limit is exceeded. True analog

control is used, with closed-loop feedback ensuring that

LT1769

1769fa

APPLICATIONS INFORMATION

WUUU

holding the battery at “float” voltage indefinitely. In this

case no external sensing of full charge is needed.

Battery Voltage Sense Resistors Selection

To minimize battery drain when the charger is off, current

through the R3/R4 divider is set at 15µA. The input current

to the OVP pin is 3nA and the error can be neglected.

With divider current set at 15µA, V

BAT

= 8.4V, R4 =

2.465/15µA = 162k and,

R3 R4 V 2.465

2.465

162k 8.4 2.465

2.465

390k

BAT

()

−

()

=−

()

Li-Ion batteries typically require float voltage accuracy of

1%. Accuracy of the LT1769 OVP voltage is ±0.5% at

25°C and ±1% over full temperature. This leads to the

possibility that very accurate (0.1%) resistors might be

needed for R3 and R4. Actually, the temperature of the

LT1769 will rarely exceed 50°C in float mode because

charging currents have tapered off to a low level, so

0.25% resistors will normally provide the required level of

overall accuracy.

When power is on, there is about 200µA of current flowing

out of the BAT and SENSE pins. If the battery is removed

during charging, and total load including R3 and R4 is less

than 200µA, V

BAT

could float up to V

even though the

loop has turned switching off. To keep V

BAT

regulated to

the battery voltage in this condition, R3 and R4 can be

chosen to draw 0.5mA and Q3 can be added to disconnect

them when power is off (Figure 4). R5 isolates the OVP pin

from any high frequency noise on V

. An alternative method

Figure 4. Disconnecting Voltage Divider

PWM

PROG

4.7k

300Ω

PROG

1µF

VN2222

LT1769

1769 F03

BAT

= (DC)(2A)

Figure 3. PWM Current Programming

12k

0.25%

4.99k

0.25%

OVP

8.4V

BAT

VN2222

LT1769

1769 F04

220k

is to use a Zener diode with a breakdown voltage two or three

volts higher than battery voltage to clamp the V

BAT

voltage.

Battery manufacturers recommend terminating the con-

stant-voltage float mode after charge current has dropped

below a specified level (typically around 10% of the full

current)

and

a further time out period of 30 to 90 minutes

has elapsed. This may extend battery life, so check with the

manufacturer for details. The circuit in Figure 5 will detect

when charge current has dropped below 270mA. This

logic signal is used to initiate a timeout period, after which

the LT1769 can be shut down by pulling the V

pin low with

an open collector or drain. Some external means must be

used to detect the need for additional charging or the

Figure 5. Current Comparator for Initiating Float Time Out

NEGATIVE EDGE

TO TIMER

1769 F04

3.3V OR 5V

ADAPTER

OUTPUT

1N4148

0.1µF

BAT

SENSE

R1*

1.6k

RS1

0.05Ω

470k

430k

560k

LT1011

1N4148

* TRIP CURRENT =

=≈ 270mA

R1(VBAT)

(R2 + R3)(RS1)

(1.6k)(8.4V)

(560k + 430k)(0.05Ω)

–

VBAT

BAT

RS3

200Ω

RS2

200Ω

LT1769

IBAT

LT1769

1769fa

APPLICATIONS INFORMATION

WUUU

is not nearly as pronounced. This makes it more difficult

to use –dV/dt as an indicator of full charge, and an

increase in battery temperature is more often used with a

temperature sensor in the battery pack. Secondly, con-

stant trickle charge may not be recommended. Instead, a

moderate level of current is used on a pulse basis (≈ 1%

to 5% duty cycle) with the time-averaged value substitut-

ing for a constant low trickle. Please contact the Linear

Technology Applications department about charge termi-

nation circuits.

If overvoltage protection is needed, R3 and R4 can be cal-

culated according to the procedure described in Lithium-

Ion Charging section. The OVP pin should be grounded if

not used.

When a microprocessor DAC output is used to control

charge current, it must be capable of sinking current at a

compliance up to 2.5V if connected directly to the PROG pin.

Thermal Calculations

If the LT1769 is used for charging currents above 1A, a

thermal calculation should be done to ensure that junction

temperature will not exceed 125°C. Power dissipation in

the IC is caused by bias and driver current, switch resis-

tance and switch transition losses. The GN package, with

a thermal resistance of 35°C/W, can provide a full 2A

charging current in many situations. A graph is shown in

the Typical Performance Characteristics section.

P 3.5mA V 1.5mA V

V7.5mA 0.012 I

55 V

PIRV

VtVI f

BIAS IN BAT

BAT 2

IN BAT

DRIVER

BAT BAT 2

SW BAT 2SW BAT

IN OL IN BAT

()()

()

()()

[]

()( )











()

()()( )

()()( )()

130

BAT

= Switch ON resistance ≈ 0.16Ω

= Effective switch overlap time ≈ 10ns

f = 200kHz

charger may be turned on periodically to complete a short

float-voltage cycle.

Current trip level is determined by the battery voltage, R1

through R3 and the sense resistor (R

). D2 generates

hysteresis in the trip level to avoid multiple comparator

transitions.

Nickel-Cadmium and Nickel-Metal-Hydride Charging

The 2A Lithium-Ion Battery Charger shown in Figure 1 can

be modified to charge NiCd or NiMH batteries. For ex-

ample, if a 2-level charge is needed; 1A when Q1 is on and

100mA when Q1 is off.

Figure 6. 2-Level Charging

5.49k

49.3k

300Ω

PROG

1µFQ1

LT1769

1769 F05

For 1A full current, the current sense resistor (R

) should

be increased to 0.1Ω so that enough signal (10mV) will be

across R

at 0.1A trickle charge to keep charging current

accurate.

For a 2-level charger, R1 and R2 are found from:

R1 2.465 2000

I R2 2.465 2000

LOW HI LOW

()()

−

All battery chargers with fast charge rates require some

means to detect full charge in the battery and terminate the

high charge current. NiCd batteries are typically charged at

high current until a temperature rise or battery voltage

decrease is detected as an indication of near full charge.

The charging current is then reduced to a much lower

value and maintained as a constant trickle charge. An

intermediate “top off” current may also be used for a fixed

time period to reduce total charge time.

NiMH batteries are similar in chemistry to NiCd but have

two differences related to charging. First, the inflection

characteristic in battery voltage as full charge is approached

LT1769

1769fa

APPLICATIONS INFORMATION

WUUU

Figure 7. Lower VBOOST

BOOST

SPIN

1769 F07

LT1769

10µF

thermal resistance of the package-board combination is

dominated by the characteristics of the board in the

immediate area of the package. This means both lateral

thermal resistance across the board and vertical thermal

resistance through the board to other copper layers. Each

layer acts as a thermal heat spreader that increases the

heat sinking effectiveness of extended areas of the board.

Total board area becomes an important factor when the

area of the board drops below about 20 square inches. The

graph in Figure 8 shows thermal resistance vs board area

for 2-layer and 4-layer boards with continuous copper

planes. Note that 4-layer boards have significantly lower

thermal resistance, but both types show a rapid increase

for reduced board areas. Figure 9 shows actual measured

lead temperatures for chargers operating at full current.

Battery voltage and input voltage will affect device power

dissipation, so the data sheet power calculations must be

used to extrapolate these readings to other situations.

Vias should be used to connect board layers together.

Planes under the charger area can be cut away from the

rest of the board and connected with vias to form both a

low thermal resistance system and to act as a ground plane

for reduced EMI.

Glue-on, chip-mounted heat sinks are effective only in

moderate power applications where the PC board copper

cannot be used, or where the board size is small. They offer

very little improvement in a properly laid out multilayer

board of reasonable size.

Higher Duty Cycle for the LT1769 Battery Charger

Maximum duty cycle for the LT1769 is typically 90%, but

this may be too low for some applications. For example, if

Example: V

= 19V, V

BAT

= 12.6V, I

BAT

= 2A:

P 3.5mA 19 1.5mA 12.6

12.6

19 7.5mA 0.012 2000mA 0.35W

2 12.6

55 19 0.43W

P2 0.16 12.6

19 10 19 2 200kHz

0.42 0.08 0.5W

BIAS

DRIVER

()()

()

()( )

[]

()( )











()

()( )( )

()()( )

=+=

−

112 6

Total Power in the IC is: 0.35 + 0.43 + 0.5 = 1.3W

Temperature rise will be (1.3W)(35°C/W) = 46°C. This

assumes that the LT1769 is properly heat sunk by con-

necting the eleven fused ground pins to expanded traces

and that the PC board has a backside or internal plane for

heat spreading.

The P

DRIVER

term can be reduced by connecting the boost

diode D2 (see Figure 7) to a lower system voltage (lower

than V

BAT

) instead of V

BAT

Then P

DRIVER

()( )()











()

IVV V

BAT BAT X X

130

For example, V

= 3.3V then:

AVV V

DRIVER

()( )( )











()

2 126 33 1 33

55 19 009

.. .

The average I

required is:

VmA

DRIVER

X==

009

33 28

The previous example shows the dramatic drop in driver

power dissipation when the boost diode (D2) is connected

to an external 3.3V source instead of the 12.6V battery.

DRIVER

drops from 0.43W to 0.09W resulting in an

approximately 12°C drop in junction temperature.

Fused-lead packages conduct most of their heat out the

leads. This makes it very important to provide as much PC

board copper around the leads as is practical. Total

LT1769

1769fa

APPLICATIONS INFORMATION

WUUU

Figure 11. Replacing the Input Diode

CHARGE CURRENT (A)

LEAD TEMPERATURE ON PINS 1, 2, 3 (°C)

1769 F09

20 0.5 11.5

NOTE: PEAK DIE TEMPERATURE

WILL BE ABOUT 15°C HIGHER AT

2A CHARGE CURRENT

VIN = 19V

VBAT = 12.3V

VBOOST = 5V

2-LAYER BOARD

ROOM TEMP = 24°C

5 IN2 BOARD

25 IN2 BOARD

BOARD AREA (IN

)

10 15 25

1769 F08

510 20 30 35

THERMAL RESISTANCE (°C/W)

MEASURED FROM AIR AMBIENT

TO DIE USING COPPER LANDS

AS SHOWN ON DATA SHEET

2-LAYER BOARD

4-LAYER BOARD

Figure 8. LT1769 Thermal Resistance Figure 10. High Duty Cycle

BOOST

SPIN

SENSE BAT

3V TO 6V C

10µF

BAT

1769 F11

0.47µF

50k

LT1769

HIGH DUTY CYCLE CONNECTION

Q1 = Si4435DY

Q2 = TP0610L

BOOST

SPIN

SENSE BAT

BAT

0.47µF

D2 LT1769

BOOST

SPIN

SENSE BAT

3V TO 6V C

10µF

BAT

1769 F10

0.47µF

D2 LT1769

STANDARD CONNECTION HIGH DUTY CYCLE CONNECTION

+ +

an 18V ±3% adapter is used to charge ten NiMH cells, the

charger must put out approximaly 15V. A total of 1.6V is

lost in the input diode, switch resistance, inductor resis-

tance and parasitics, so the required duty cycle is

15/16.4 = 91.4%. The duty cycle can be extended to 93%

by restricting boost voltage to 5V instead of using V

BAT

is normally done. This lower boost voltage also reduces

power dissipation in the LT1769, so it is a win-win

decision. Connect an external source of 3V to 6V at V

node in Figure 10 with a 10µF C

bypass capacitor.

Lower Dropout Voltage

For even lower dropout and/or reducing heat on the board,

the input diode D3 can be replaced with a FET (see Figure

11). Connect a P-channel FET in place of the input diode

with its gate connected to the battery causing the FET to

turn off when the input voltage goes low. The problem is

that the gate must be pumped low so that the FET is fully

turned on even when the input is only a volt or two above

the battery voltage. Also there is a turn-off speed issue.

The FET should turn off instantly when the input is dead

shorted to avoid large current surges from the battery

back through the charger into the FET. Gate capacitance

slows turn-off, so a small P-channel (Q2) is added to

discharge the gate capacitance quickly in the event of an

input short. The Q2 body diode creates the necessary

pumping action to keep the gate of Q1 low during normal

operation. Note that Q1 and Q2 have a V

spec limit of

20V. This restricts V

to a maximum of 20V. For low

dropout operation with V

> 20V consult factory.

Figure 9. LT1769 Lead Temperature

LT1769

1769fa

APPLICATIONS INFORMATION

WUUU

Optional Diode Connections

The typical application in Figure 1 shows a single diode

(D3) to isolate the V

pin from the adaptor input and to

block reverse input voltage (both steady state and tran-

sient). This simple connection may be unacceptable in

situations where the system load must be powered from

the battery when the adapter input power is removed. As

shown in Figure 12, a parasitic diode exists from the SW

pin to the V

pin in the LT1769. When the input power is

removed, this diode will become forward biased and will

provide a current path from the battery to the system load.

Because of diode power limitations, it is not recommended

to power the system load through the internal parasitic

diode. To safely power the system load from the battery,

an additional Schottky diode (D4) is needed. For minimum

losses, D4 could be replaced by a low R

DS(ON)

MOSFET

which is turned on when the adapter power is removed.

Layout Considerations

Switch rise and fall times are under 10ns for maximum

efficiency. To minimize radiation, the catch diode, SW pin

and input bypass capacitor leads should be kept as short

as possible. A ground plane should be used under the

switching circuitry to prevent interplane coupling and to

act as a thermal spreading path. All ground pins should be

connected to expanded traces for low thermal resistance.

The fast-switching high current ground path, including the

switch, catch diode and input capacitor, should be kept

very short. Catch diode and input capacitor should be

close to the chip and terminated to the same point. This

path contains nanosecond rise and fall times with several

amps of current. The other paths contain only DC and/or

200kHz tri-wave and are less critical. Figure 13 indicates

the high speed, high current switching path. Figure 14

shows critical path layout. Contact Linear Technology for

the LT1769 circuit PCB layout or Gerber file.

CLP

CLN

ADAPTER

SYSTEM

LOAD

500Ω

1µF

LT1769

INTERNAL

PARASITIC

DIODE

1769 F12a

Figure 12. Modified Diode Connection Figure 13. High Speed Switching Path

1769 F13

BAT

HIGH

FREQUENCY

CIRCULATING

PATH

BAT

SWITCH NODE

OUT

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-

tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

OUT

GND

1769 F14

GND

NOTE: CONNECT ALL GND PINS TO EXPANDED PC LANDS FOR PROPER HEAT SINKING

GND

BOOST

GND

OVP

CLP

CLN

COMP1

SENSE

GND

CC1

CC2

CC3

GND

PROG

OUT

COMP2

BAT

SPIN

GND

Figure 14. Critical Electrical and Thermal Path Layout

LT1769

1769fa

LT/TP 1101 1.5K REV A • PRINTED IN USA

 LINEAR TECHNOLOGY CORPORATION 1999

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

(408) 432-1900

●

FAX: (408) 434-0507

●

www.linear.com

PART NUMBER DESCRIPTION COMMENTS

LT1372/LT1377 500kHz/1MHz Step-Up Switching Regulators High Frequency, Small Inductor, High Efficiency Switchers, 1.5A Switch

LT1376 500kHz Step-Down Switching Regulator High Frequency, Small Inductor, High Efficiency Switcher, 1.5A Switch

LT1505 High Current, High Efficiency Battery Charger 94% Efficiency, Synchronous Current Mode PWM

LT1510 Constant-Voltage/Constant-Current Battery Charger Up to 1.5A Charge Current for Lithium-Ion, NiCd and NiMH Batteries

LT1511 Constant-Voltage/Constant-Current Battery Charger Up to 3A Charge Current for Lithium-Ion, NiCd and NiMH Batteries

LT1512/LT1513 SEPIC Battery Chargers V

Can Be Higher or Lower Than Battery Voltage

LTC1729 Li-Ion Battery Charger Termination Controller Preconditioning If Cell < 2.7V, 3hr Time-Out, C/10 Detection, Temp Sensor

Pin, Charger and Battery Detection

LTC1759 SMBus Smart Battery Charger 94% Efficiency with Input Current Limiting, Up to 8A I

CHG

LTC1960 Dual Battery Charger and Selector with SPI Interface I

CHARGE

Up to 6A, Fast Charge, Longer Battery Life, Crisis Management

RELATED PARTS

PACKAGE DESCRIPTION

GN Package

28-Lead Plastic SSOP (Narrow .150 Inch)

(Reference LTC DWG # 05-08-1641)

GN28 (SSOP) 1098

* DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH

SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD

FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

0.016 – 0.050

(0.406 – 1.270)

0.015 ± 0.004

(0.38 ± 0.10) × 45°

0° – 8° TYP

0.0075 – 0.0098

(0.191 – 0.249)

0.053 – 0.069

(1.351 – 1.748)

0.008 – 0.012

(0.203 – 0.305)

0.004 – 0.009

(0.102 – 0.249)

0.0250

(0.635)

BSC

0.386 – 0.393*

(9.804 – 9.982)

345678 9 10 11 12

0.229 – 0.244

(5.817 – 6.198)

0.150 – 0.157**

(3.810 – 3.988)

202122232425262728 19 18 17

13 14

1615

0.033

(0.838)

REF

FE Package

20-Lead Plastic TSSOP (4.4mm)

(Reference LTC DWG # 05-08-1663)

Exposed Pad Variation CB

FE20 (CB) TSSOP 0203

0.09 – 0.20

(.0036 – .0079)

0° – 8°

RECOMMENDED SOLDER PAD LAYOUT

0.45 – 0.75

(.018 – .030)

4.30 – 4.50*

(.169 – .177)

1.20

(.047)

MAX

0.05 – 0.15

(.002 – .006)

0.65

(.0256)

BSC 0.195 – 0.30

(.0077 – .0118)

2.74

(.108)

0.45 ±0.05

0.65 BSC

4.50 ±0.10

6.60 ±0.10

1.05 ±0.10

3.86

(.152)

6.40

BSC

134

5678910

111214 13

6.40 – 6.60*

(.252 – .260)

3.86

(.152)

2.74

(.108)

20 1918 17 16 15

MILLIMETERS

(INCHES) *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH

SHALL NOT EXCEED 0.150mm (.006") PER SIDE

NOTE:

1. CONTROLLING DIMENSION: MILLIMETERS

2. DIMENSIONS ARE IN

3. DRAWING NOT TO SCALE

SEE NOTE 4

4. RECOMMENDED MINIMUM PCB METAL SIZE

FOR EXPOSED PAD ATTACHMENT