LT3573
1
3573fc
Features
applications
Description
Isolated Flyback Converter
without an Opto-Coupler
The LT
®
3573 is a monolithic switching regulator specifi-
cally designed for the isolated flyback topology. No third
winding or opto-isolator is required for regulation. The
part senses the isolated output voltage directly from the
primary-side flyback waveform. A 1.25A, 60V NPN power
switch is integrated along with all control logic into a
16-lead MSOP package.
The LT3573 operates with input supply voltages from
3V to 40V, and can deliver output power up to 7W with
no external power devices.The LT3573 utilizes boundary
mode operation to provide a small magnetic solution with
improved load regulation.
5V Isolated Flyback Converter
n 3V to 40V Input Voltage Range
n 1.25A, 60V Integrated NPN Power Switch
n Boundary Mode Operation
n No Transformer Third Winding or
Opto-Isolator Required for Regulation
n Improved Primary-Side Winding Feedback
Load Regulation
n VOUT Set with Two External Resistors
n BIAS Pin for Internal Bias Supply and Power
NPN Driver
n Programmable Soft-Start
n Programmable Power Switch Current Limit
n Thermally Enhanced 16-Lead MSOP
n Industrial, Automotive and Medical Isolated
Power Supplies
Load Regulation
SHDN/UVLO
TC
RILIM
SS
RFB
RREF
SW
VC GND TEST BIAS
LT3573
3573 TA01
28.7k 10k 20k
VIN
12V TO 24V
VOUT+
5V, 0.7A
VOUT–
VIN
3:1
357k
51.1k
10µF
2.6µH24µH47µF
10nF 1nF 4.7µF
6.04k
2k
80.6k
B340A
PMEG6010
0.22µF
IOUT (mA)
0
OUTPUT VOLTAGE ERROR (%)
1
2
0
–1
400 800
200 600 1000 1200 1400
–2
–3
3
3573 TA01b
VIN = 12V
VIN = 24V
typical application
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks
and No RSENSE and ThinSOT are trademarks of Linear Technology Corporation. All other
trademarks are the property of their respective owners. Protected by U.S. Patents, including
5438499 and 7471522.
LT3573
2
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absolute MaxiMuM ratings
SW ............................................................................60V
VIN, SHDN/UVLO, RFB, BIAS .....................................40V
SS, VC, TC, RREF, RILIM ...............................................5V
Maximum Junction Temperature .......................... 125°C
Operating Junction Temperature Range (Note 2)
LT3573E ............................................–40°C to 125°C
Storage Temperature Range ..................–65°C to 150°C
orDer inForMation
electrical characteristics
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, unless otherwise noted.
LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LT3573EMSE#PBF LT3573EMSE#TRPBF 3573 16-Lead Plastic MSOP –40°C to 125°C
LT3573IMSE#PBF LT3573IMSE#TRPBF 3573 16-Lead Plastic MSOP –40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
PARAMETER CONDITIONS MIN TYP MAX UNITS
Input Voltage Range l3 40 V
Quiescent Current SS = 0V
VSHDN/UVLO = 0V 3.5
0
1mA
µA
Soft-Start Current SS = 0.4V 7 µA
SHDN/UVLO Pin Threshold UVLO Pin Voltage Rising l1.15 1.22 1.29 V
SHDN/UVLO Pin Hysteresis Current VUVLO = 1V 2 2.5 3 µA
Soft-Start Threshold 0.7 V
Maximum Switching Frequency 1000 kHz
Switch Current Limit RILIM = 10k 1.25 1.55 1.85 A
Minimum Current Limit VC = 0V 200 mA
Switch VCESAT ISW = 0.5A 150 250 mV
RREF Voltage VIN = 3V
l
1.21
1.20 1.23 1.25
1.25 V
RREF Voltage Line Regulation 3V < VIN < 40V 0.01 0.03 %/ V
RREF Pin Bias Current (Note 3) l100 600 nA
IREF Reference Current Measured at RFB Pin with RREF = 6.49k 190 µA
pin conFiguration
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
TOP VIEW
MSE PACKAGE
16-LEAD PLASTIC MSOP
GND
TEST
GND
SW
VIN
BIAS
SHDN/UVLO
GND
GND
TC
RREF
RFB
VC
RILIM
SS
GND
17
TJMAX = 125°C, θJA = 50°C/W, θJC = 10°C/W
EXPOSED PAD (PIN 17) IS GND, MUST BE CONNECTED TO GND
LT3573
3
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Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3573E is guaranteed to meet performance specifications
from 0°C to 125°C junction temperature. Specifications over the –40°C
to 125°C operating junction temperature range are assured by design
characterization and correlation with statistical process controls. The
LT3573I is guaranteed over the full –40°C to 125°C operating junction
temperature range.
Note 3: Current flows out of the RREF pin.
PARAMETER CONDITIONS MIN TYP MAX UNITS
Error Amplifier Voltage Gain VIN = 3V 150 V/V
Error Amplifier Transconductance DI = 10µA, VIN = 3V 150 µmhos
Minimum Switching Frequency VC = 0.35V 40 kHz
TC Current into RREF RTC = 20.1k 27.5 µA
BIAS Pin Voltage IBIAS = 30mA 2.9 3 3.1 V
electrical characteristics
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, unless otherwise noted.
typical perForMance characteristics
Output Voltage Quiescent Current Bias Pin Voltage
TA = 25°C, unless otherwise noted.
TEMPERATURE (°C)
–50
4.80
VOUT (V)
4.85
4.95
5.00
5.05
5.20
5.15
050 75
4.90
5.10
–25 25 100 125
3573 G01
TEMPERATURE (°C)
–50
0
IQ (mA)
2
3
4
7
6
050 75
1
5
–25 25 100 125
3573 G02
VIN = 40V
VIN = 5V
TEMPERATURE (°C)
–50
2.0
BIAS VOLTAGE (V)
2.4
2.6
2.8
3.2
050 75
2.2
3.0
–25 25 100 125
3573 G03
VIN = 40V
VIN = 12V
LT3573
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typical perForMance characteristics
Switch VCESAT Switch Current Limit Switch Current Limit vs RILIM
SHDN/UVLO Falling Threshold SS Pin Current
TA = 25°C, unless otherwise noted.
SWITCH CURRENT (A)
0
0
SWITCH VCESAT VOLTAGE (mV)
100
200
300
0.25 0.50 1.000.75 1.25
400
50
150
250
350
1.50
3573 G04
125°C
25°C
–50°C
TEMPERATURE (°C)
–50
CURRENT LIMIT (A)
1.2
1.4
1.6
1.0
0.8
–25 250 50 75 100 125
0.2
0
0.6
1.8
0.4
3573 G05
MAXIMUM CURRENT LIMIT
MINIMUM CURRENT LIMIT
RILIM = 10k
TEMPERATURE (°C)
–60
SHDN/UVLO VOLTAGE (V)
1.24
1.26
80
1.22
–20 20
–40 120
0 40 100
60 140
1.20
1.18
1.28
3573 G07
TEMPERATURE (°C)
–60
SS PIN CURRENT (µA)
8
10
80
6
4
–20 20
–40 120
0 40 100
60 140
2
0
12
3573 G08
LT3573
5
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pin Functions
GND: Ground.
TEST: This pin is used for testing purposes only and must
be connected to ground for the part to operate properly.
SW: Collector Node of the Output Switch. This pin has
large currents flowing through it. Keep the traces to the
switching components as short as possible to minimize
electromagnetic radiation and voltage spikes.
VIN: Input Voltage. This pin supplies current to the internal
start-up circuitry and as a reference voltage for the DCM
comparator and feedback circuitry. This pin must be locally
bypassed with a capacitor.
BIAS: Bias Voltage. This pin supplies current to the switch
driver and internal circuitry of the LT3573. This pin must
be locally bypassed with a capacitor. This pin may also be
connected to VIN if a third winding is not used and if VIN
≤ 15V. If a third winding is used, the BIAS voltage should
be lower than the input voltage for proper operation.
SHDN/UVLO: Shutdown/Undervoltage Lockout. A resistor
divider connected to VIN is tied to this pin to program the
minimum input voltage at which the LT3573 will operate.
At a voltage below ~0.7V, the part draws no quiescent
current. When below 1.25V and above ~0.7V, the part will
draw 10µA of current, but internal circuitry will remain off.
Above 1.25V, the internal circuitry will start and a 10µA
current will be fed into the SS pin. When this pin falls
below 1.25V, 2.5µA will be pulled from the pin to provide
programmable hysteresis for UVLO.
RILIM: Maximum Current Limit Adjust Pin. A resistor
should be tied to this pin to ground to set the current
limit. Use a 10k resistor for the full current capabilities
of the switch.
SS: Soft-Start Pin. Place a soft-start capacitor here to
limit start-up inrush current and output voltage ramp
rate. Switching starts when the voltage at this pin reaches
~0.7V.
VC: Compensation Pin for Internal Error Amplifier. Connect
a series RC from this pin to ground to compensate the
switching regulator. A 100pF capacitor in parallel helps
eliminate noise.
RFB: Input Pin for External Feedback Resistor. This pin is
connected to the transformer primary (VSW). The ratio
of this resistor to the RREF resistor, times the internal
bandgap reference, determines the output voltage (plus
the effect of any non-unity transformer turns ratio). The
average current through this resistor during the flyback
period should be approximately 200µA. For nonisolated
applications, this pin should be connected to VIN.
RREF: Input Pin for External Ground-Referred Reference
Resistor. This resistor should be in the range of 6k, but
for convenience, need not be precisely this value. For
nonisolated applications, a traditional resistor voltage
divider may be connected to this pin.
TC: Output Voltage Temperature Compensation. Connect
a resistor to ground to produce a current proportional to
absolute temperature to be sourced into the RREF node.
ITC = 0.55V/RTC.
Exposed Pad: Ground. The Exposed Pad of the package
provides both electrical contact to ground and good thermal
contact to the printed circuit board. The Exposed Pad must
be soldered to the circuit board for proper operation and
should be well connected with many vias to an internal
ground plane.
LT3573
6
3573fc
block DiagraM
FLYBACK
ERROR
AMP
MASTER
LATCH
CURRENT
COMPARATOR
BIAS
R1
R2
C3
R6
VOUT+
VOUT–
VIN
TC
BIAS
SS
SWVIN
VIN
GND
V1
120mV
1.22V
VC
D1
T1
N:1
I1
7µA
I2
20µA
RSENSE
0.02Ω
C2
C1 L1A L1B
R3
R4
C5
+
–
INTERNAL
REFERENCE
AND
REGULATORS
OSCILLATOR
TC
CURRENT ONE
SHOT
RQ
S
S
gm
+
–A1
+
–A5
+
–
+
–
A2
A4
2.5µA
+
–
3573 BD
Q2
R7
C4
R5
Q3
1.22V
Q4
Q1
DRIVER
SHDN/UVLO
RILIM
RFB
RREF
LT3573
7
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operation
The LT3573 is a current mode switching regulator IC
designed specifically for the isolated flyback topology.
The special problem normally encountered in such cir-
cuits is that information relating to the output voltage on
the isolated secondary side of the transformer must be
communicated to the primary side in order to maintain
regulation. Historically, this has been done with opto-
isolators or extra transformer windings. Opto-isolator
circuits waste output power and the extra components
increase the cost and physical size of the power supply.
Opto-isolators can also exhibit trouble due to limited
dynamic response, nonlinearity, unit-to-unit variation
and aging over life. Circuits employing extra transformer
windings also exhibit deficiencies. Using an extra wind-
ing adds to the transformer’s physical size and cost, and
dynamic response is often mediocre.
The LT3573 derives its information about the isolated
output voltage by examining the primary-side flyback
pulse waveform. In this manner, no opto-isolator nor extra
transformer winding is required for regulation. The output
voltage is easily programmed with two resistors. Since this
IC operates in boundary control mode, the output voltage
is calculated from the switch pin when the secondary cur-
rent is almost zero. This method improves load regulation
without external resistors and capacitors.
The Block Diagram shows an overall view of the system.
Many of the blocks are similar to those found in traditional
switching regulators including: internal bias regulator,
oscillator, logic, current amplifier and comparator, driver,
and output switch. The novel sections include a special
flyback error amplifier and a temperature compensation
circuit. In addition, the logic system contains additional
logic for boundary mode operation, and the sampling
error amplifier.
The LT3573 features a boundary mode control method,
where the part operates at the boundary between continu-
ous conduction mode and discontinuous conduction mode.
The VC pin controls the current level just as it does in normal
current mode operation, but instead of turning the switch
on at the start of the oscillator period, the part detects
when the secondary side winding current is zero.
Boundary Mode Operation
Boundary mode is a variable frequency, current-mode
switching scheme. The switch turns on and the inductor
current increases until a VC pin controlled current limit. The
voltage on the SW pin rises to the output voltage divided
by the secondary-to-primary transformer turns ratio plus
the input voltage. When the secondary current through
the diode falls to zero, the SW pin voltage falls below VIN.
A discontinuous conduction mode (DCM) comparator
detects this event and turns the switch back on.
Boundary mode returns the secondary current to zero
every cycle, so the parasitic resistive voltage drops do not
cause load regulation errors. Boundary mode also allows
the use of a smaller transformer compared to continuous
conduction mode and no subharmonic oscillation.
At low output currents the LT3573 delays turning on the
switch, and thus operates in discontinuous mode. Unlike
a traditional flyback converter, the switch has to turn on
to update the output voltage information. Below 0.6V on
the VC pin, the current comparator level decreases to
its minimum value, and the internal oscillator frequency
decreases in frequency. With the decrease of the internal
oscillator, the part starts to operate in DCM. The output
current is able to decrease while still allowing a minimum
switch off-time for the error amp sampling circuitry. The
typical minimum internal oscillator frequency with VC
equal to 0V is 40kHz.
LT3573
8
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ERROR AMPLIFIER—PSEUDO DC THEORY
In the Block Diagram, the RREF (R4) and RFB (R3) resistors
can be found. They are external resistors used to program
the output voltage. The LT3573 operates much the same
way as traditional current mode switchers, the major
difference being a different type of error amplifier which
derives its feedback information from the flyback pulse.
Operation is as follows: when the output switch, Q1, turns
off, its collector voltage rises above the VIN rail. The am-
plitude of this flyback pulse, i.e., the difference between
it and VIN, is given as:
VFLBK = (VOUT + VF + ISEC • ESR) • NPS
VF = D1 forward voltage
ISEC = Transformer secondary current
ESR = Total impedance of secondary circuit
NPS = Transformer effective primary-to-secondary
turns ratio
The flyback voltage is then converted to a current by
the action of RFB and Q2. Nearly all of this current flows
through resistor RREF to form a ground-referred volt-
age. This voltage is fed into the flyback error amplifier.
The flyback error amplifier samples this output voltage
information when the secondary side winding current is
zero. The error amplifier uses a bandgap voltage, 1.23V,
as the reference voltage.
The relatively high gain in the overall loop will then cause
the voltage at the RREF resistor to be nearly equal to the
bandgap reference voltage VBG. The relationship between
VFLBK and VBG may then be expressed as:
aV
R
V
Ror
V V R
R
FLBK
FB
BG
REF
FLBK BG FB
REF
=
=
,
1
a
a = Ratio of Q1 IC to IE, typically
≈
0.986
VBG = Internal bandgap reference
In combination with the previous VFLBK expression yields
an expression for VOUT, in terms of the internal reference,
programming resistors, transformer turns ratio and diode
forward voltage drop:
V V R
R N V I ES
OUT BG FB
REF PS F SEC
=
− −
1
a( RR)
Additionally, it includes the effect of nonzero secondary
output impedance (ESR). This term can be assumed to
be zero in boundary control mode. More details will be
discussed in the next section.
Temperature Compensation
The first term in the VOUT equation does not have a tem-
perature dependence, but the diode forward drop has a
significant negative temperature coefficient. To compen-
sate for this, a positive temperature coefficient current
source is connected to the RREF pin. The current is set by
a resistor to ground connected to the TC pin. To cancel the
temperature coefficient, the following equation is used:
d
d
d
d
d
V
T
R
R N
V
Tor
RR
N V
F FB
TC PS
TC
TC FB
PS
= −
=−
• • ,
•
1
1
FF
TC FB
PS
T
V
T
R
N/ •
d
d
d≈
(dVF/dT) = Diode’s forward voltage temperature
coefficient
(dVTC/dT) = 2mV
VTC = 0.55V
The resistor value given by this equation should also be
verified experimentally, and adjusted if necessary to achieve
optimal regulation over temperature.
The revised output voltage is as follows:
V V R
R N V
V
R
OUT BG FB
REF PS F
TC
TC
=
−
−
1
a
• – ( )
R
NI ESR
FB
PS SEC
a
applications inForMation
LT3573
9
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applications inForMation
ERROR AMPLIFIER—DYNAMIC THEORY
Due to the sampling nature of the feedback loop, there
are several timing signals and other constraints that are
required for proper LT3573 operation.
Minimum Current Limit
The LT3573 obtains output voltage information from the
SW pin when the secondary winding conducts current.
The sampling circuitry needs a minimum amount of time
to sample the output voltage. To guarantee enough time,
a minimum inductance value must be maintained. The
primary-side magnetizing inductance must be chosen
above the following value:
L V t
IN V N µH
V
PRI OUT MIN
MIN PS OUT PS
≥ =
• • • • .1 4
tMIN = minimum off-time, 350ns
IMIN = minimum current limit, 250mA
The minimum current limit is higher than that on the Elec-
trical Characteristics table due to the overshoot caused by
the comparator delay.
Leakage Inductance Blanking
When the output switch first turns off, the flyback pulse
appears. However, it takes a finite time until the transformer
primary-side voltage waveform approximately represents
the output voltage. This is partly due to the rise time on
the SW node, but more importantly due to the trans-
former leakage inductance. The latter causes a very fast
voltage spike on the primary-side of the transformer that
is not directly related to output voltage (some time is also
required for internal settling of the feedback amplifier
circuitry). The leakage inductance spike is largest when
the power switch current is highest.
In order to maintain immunity to these phenomena, a fixed
delay is introduced between the switch turn-off command
and the beginning of the sampling. The blanking is internally
set to 150ns. In certain cases, the leakage inductance may
not be settled by the end of the blanking period, but will
not significantly affect output regulation.
Selecting RFB and RREF Resistor Values
The expression for VOUT, developed in the Operation sec-
tion, can be rearranged to yield the following expression
for RFB:
RR N V V V
V
FB
REF PS OUT F TC
BG
=+
( )
+
•a
where,
VOUT = Output voltage
VF = Switching diode forward voltage
a = Ratio of Q1, IC to IE, typically 0.986
NPS = Effective primary-to-secondary turns ratio
VTC = 0.55V
The equation assumes the temperature coefficients of
the diode and VTC are equal, which is a good first-order
approximation.
Strictly speaking, the above equation defines RFB not as an
absolute value, but as a ratio of RREF. So, the next ques-
tion is, “What is the proper value for RREF?” The answer
is that RREF should be approximately 6.04k. The LT3573
is trimmed and specified using this value of RREF. If the
impedance of RREF varies considerably from 6.04k, ad-
ditional errors will result. However, a variation in RREF of
several percent is acceptable. This yields a bit of freedom
in selecting standard 1% resistor values to yield nominal
RFB/RREF ratios.
Tables 1-4 are useful for selecting the resistor values for
RREF and RFB with no equations. The tables provide RFB,
RREF and RTC values for common output voltages and
common winding ratios.
Table 1. Common Resistor Values for 1:1 Transformers
VOUT (V) NPS RFB (kΩ) RREF (kΩ) RTC (kΩ)
3.3 1.00 18.7 6.04 19.1
5 1.00 27.4 6.04 28
12 1.00 64.9 6.04 66.5
15 1.00 80.6 6.04 80.6
20 1.00 107 6.04 105
LT3573
10
3573fc
applications inForMation
Table 2. Common Resistor Values for 2:1 Transformers
VOUT (V) NPS RFB (kΩ) RREF (kΩ) RTC (kΩ)
3.3 2.00 37.4 6.04 18.7
5 2.00 56 6.04 28
12 2.00 130 6.04 66.5
15 2.00 162 6.04 80.6
Table 3. Common Resistor Values for 3:1 Transformers
VOUT (V) NPS RFB (kΩ) RREF (kΩ) RTC (kΩ)
3.3 3.00 56.2 6.04 20
5 3.00 80.6 6.04 28.7
10 3.00 165 6.04 54.9
Table 4. Common Resistor Values for 4:1 Transformers
VOUT (V) NPS RFB (kΩ) RREF (kΩ) RTC (kΩ)
3.3 4.00 76.8 6.04 19.1
5 4.00 113 6.04 28
Output Power
A flyback converter has a complicated relationship be-
tween the input and output current compared to a buck
or a boost. A boost has a relatively constant maximum
input current regardless of input voltage and a buck has a
relatively constant maximum output current regardless of
input voltage. This is due to the continuous nonswitching
behavior of the two currents. A flyback converter has both
discontinuous input and output currents which makes it
similar to a nonisolated buck-boost. The duty cycle will
affect the input and output currents, making it hard to
predict output power. In addition, the winding ratio can
be changed to multiply the output current at the expense
of a higher switch voltage.
The graphs in Figures 1-3 show the maximum output
power possible for the output voltages 3.3V, 5V, and 12V.
The maximum power output curve is the calculated output
power if the switch voltage is 50V during the off-time. To
achieve this power level at a given input, a winding ratio
value must be calculated to stress the switch to 50V,
resulting in some odd ratio values. The curves below are
examples of common winding ratio values and the amount
of output power at given input voltages.
One design example would be a 5V output converter with
a minimum input voltage of 20V and a maximum input
voltage of 30V. A three-to-one winding ratio fits this design
example perfectly and outputs close to six watts at 30V
but lowers to five watts at 20V.
Figure 1. Output Power for 3.3V Output Figure 2. Output Power for 5V Output Figure 3. Output Power for 12V Output
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
6
7
35
5
4
10 20
515 25 30 40
1
0
3
8
2
3573 F01
N = 3:1
N = 7:1
N = 4:1
N = 10:1
MAXIMUM
OUTPUT
POWER
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
6
7
35
5
4
10 20
515 25 40
30 45
1
0
3
8
2
3573 F02
N = 7:1
N = 5:1
N = 2:1
N = 3:1
MAXIMUM
OUTPUT
POWER
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
6
7
35
5
4
10 20
515 25 40
30 45
1
0
3
8
2
3573 F03
N = 3:1
N = 2:1
N = 1:1
MAXIMUM
OUTPUT
POWER
LT3573
11
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applications inForMation
TRANSFORMER DESIGN CONSIDERATIONS
Transformer specification and design is perhaps the most
critical part of successfully applying the LT3573. In addition
to the usual list of caveats dealing with high frequency
isolated power supply transformer design, the following
information should be carefully considered.
Linear Technology has worked with several leading mag-
netic component manufacturers to produce pre-designed
flyback transformers for use with the LT3573. Table 5 shows
the details of several of these transformers.
Table 5. Predesigned Transformers—Typical Specifications, Unless Otherwise Noted
TRANSFORMER
PART NUMBER
SIZE (W × L × H)
(mm)
LPRI
(µH)
LLEAKAGE
(nH) NP:NS:NB
RPRI
(mΩ)
RSEC
(mΩ) VENDOR
TARGET
APPLICATIONS
PA2362NL 15.24 × 13.1 × 11.45 24 550 4:1:1 117 9.5 Pulse Engineering 24V→3.3V, 1.5A
PA2454NL 15.24 × 13.1 × 11.45 24 430 3:1:1 82 11 Pulse Engineering 24V→5V, 1A
PA2455NL 15.24 × 13.1 × 11.45 25 450 2:1:1 82 22 Pulse Engineering 24V→12V, 0.5A
PA2456NL 15.24 × 13.1 × 11.45 25 390 1:1:1 82 84 Pulse Engineering 12V→12V, 0.3A
24V→12V, 0.4A
36V→5V, 0.6A
PA2617NL 12.70 × 10.67 × 9.14 21 245 1:1:0.33 164 166 Pulse Engineering 24V→15V, 0.4A
PA2626NL 12.70 × 10.67 × 9.14 30 403 3:1:1 240 66 Pulse Engineering 24V→5V, 1A
PA2627NL 15.24 × 13.1 × 11.45 50 766 3:1:1 420 44 Pulse Engineering 24V→5V, 1A
GA3429-BL 15.24 × 12.7 × 11.43 25 566 4:1:1 95 7.5 Coilcraft 24V→3.3V, 1.5A
750310471 15.24 × 13.3 × 11.43 25 350 3:1:1 57 11 Würth Elektronik 24V→5V, 1A
750310559 15.24 × 13.3 × 11.43 24 400 4:1:1 51 16 Würth Elektronik 24V→3.3V, 1.5A
750310562 15.24 × 13.3 × 11.43 25 330 2:1:1 60 20 Würth Elektronik 24V→12V, 0.5A
750310563 15.24 × 13.3 × 11.43 25 325 1:1:0.5 60 60 Würth Elektronik 12V→12V, 0.3A
24V→12V, 0.4A
36V→5V, 0.6A
750310564 15.24 × 13.3 × 11.43 63 450 3:1:1 115 50 Würth Elektronik 24V→±5V, 0.5A
750310799 9.14 × 9.78 × 10.54 25 125 1:1:0.33 60 74 Würth Elektronik 24V→15V, 0.4A
750370040 9.14 × 9.78 × 10.54 30 150 3:1:1 60 12.5 Würth Elektronik 24V→5V, 1A
750370041 9.14 × 9.78 × 10.54 50 450 3:1:1 190 26 Würth Elektronik 24V→5V, 1A
750370047 13.35 × 10.8 × 9.14 30 150 3:1:1 60 12.5 Würth Elektronik 24V→5V, 1A
750311681 17.75 × 13.46 × 12.70 100 3000 1:10 220 28500 Würth Elektronik 12V→300V, 5mA
L11-0059 9.52 × 9.52 × 4.95 24 3:1 266 266 BH Electronics 24V→5V, 1A
L10-1019 9.52 × 9.52 × 4.95 18 1:1 90 90 BH Electronics 5V→5V, 0.2A
LT3573
12
3573fc
applications inForMation
Turns Ratio
Note that when using an RFB/RREF resistor ratio to set
output voltage, the user has relative freedom in selecting
a transformer turns ratio to suit a given application. In
contrast, simpler ratios of small integers, e.g., 1:1, 2:1,
3:2, etc., can be employed to provide more freedom in
setting total turns and mutual inductance.
Typically, the transformer turns ratio is chosen to maximize
available output power. For low output voltages (3.3V or 5V),
a N:1 turns ratio can be used with multiple primary windings
relative to the secondary to maximize the transformer’s
current gain (and output power). However, remember that
the SW pin sees a voltage that is equal to the maximum
input supply voltage plus the output voltage multiplied by
the turns ratio. This quantity needs to remain below the
ABS MAX rating of the SW pin to prevent breakdown of
the internal power switch. Together these conditions place
an upper limit on the turns ratio, N, for a given application.
Choose a turns ratio low enough to ensure:
NV V
V V
IN MAX
OUT F
<+
50 – ( )
For larger N:1 values, a transformer with a larger physical
size is needed to deliver additional current and provide a
large enough inductance value to ensure that the off-time is
long enough to accurately measure the output voltage.
For lower output power levels, a 1:1 or 1:N transformer
can be chosen for the absolute smallest transformer size.
A 1:N transformer will minimize the magnetizing induc-
tance (and minimize size), but will also limit the available
output power. A higher 1:N turns ratio makes it possible
to have very high output voltages without exceeding the
breakdown voltage of the internal power switch.
Linear Technology has worked with several magnetic
component manufacturers to produce predesigned flyback
transformers for use with the LT3573. Table 5 shows the
details of several of these transformers.
Leakage Inductance
Transformer leakage inductance (on either the primary or
secondary) causes a voltage spike to appear at the primary
after the output switch turns off. This spike is increasingly
prominent at higher load currents where more stored
energy must be dissipated. In most cases, a snubber
circuit will be required to avoid overvoltage breakdown at
the output switch node. Transformer leakage inductance
should be minimized.
An RCD (resistor capacitor diode) clamp, shown in
Figure 4, is required for most designs to prevent the
leakage inductance spike from exceeding the breakdown
voltage of the power device. The flyback waveform is
depicted in Figure 5. In most applications, there will be a
very fast voltage spike caused by a slow clamp diode that
may not exceed 60V. Once the diode clamps, the leakage
inductance current is absorbed by the clamp capacitor.
This period should not last longer than 150ns so as not to
interfere with the output regulation, and the voltage during
this clamp period must not exceed 55V. The clamp diode
turns off after the leakage inductance energy is absorbed
and the switch voltage is then equal to:
VSW(MAX) = VIN(MAX) + N(VOUT + VF)
This voltage must not exceed 50V. This same equation
also determines the maximum turns ratio.
When choosing the snubber network diode, careful atten-
tion must be paid to maximum voltage seen by the SW
pin. Schottky diodes are typically the best choice to be
used in the snubber, but some PN diodes can be used if
they turn on fast enough to limit the leakage inductance
spike. The leakage spike must always be kept below 60V.
Figures 6 and 7 show the SW pin waveform for a 24VIN,
5VOUT application at a 1A load current. Notice that the
leakage spike is very high (more than 65V) with the “bad”
diode, while the “good” diode effectively limits the spike
to less than 55V.
LT3573
13
3573fc
Figure 5. Maximum Voltages for SW Pin Flyback WaveformFigure 4. RCD Clamp
Figure 6. Good Snubber Diode Limits SW Pin Voltage Figure 7. Bad Snubber Diode Does Not Limit SW Pin Voltage
< 50V
< 55V
< 60V
VSW
tOFF > 350ns
TIME
tSP < 150ns 3573 F05
100ns/DIV
10V/DIV
3573 F06
100ns/DIV
10V/DIV
3573 F07
3573 F04
LS
D
R
+
–C
applications inForMation
LT3573
14
3573fc
Secondary Leakage Inductance
In addition to the previously described effects of leakage
inductance in general, leakage inductance on the second-
ary in particular exhibits an additional phenomenon. It
forms an inductive divider on the transformer secondary
that effectively reduces the size of the primary-referred
flyback pulse used for feedback. This will increase the
output voltage target by a similar percentage. Note that
unlike leakage spike behavior, this phenomenon is load
independent. To the extent that the secondary leakage
inductance is a constant percentage of mutual inductance
(over manufacturing variations), this can be accommodated
by adjusting the RFB/RREF resistor ratio.
Winding Resistance Effects
Resistance in either the primary or secondary will reduce
overall efficiency (POUT/PIN). Good output voltage regula-
tion will be maintained independent of winding resistance
due to the boundary mode operation of the LT3573.
Bifilar Winding
A bifilar, or similar winding technique, is a good way to
minimize troublesome leakage inductances. However, re-
member that this will also increase primary-to-secondary
capacitance and limit the primary-to-secondary breakdown
voltage, so, bifilar winding is not always practical. The
Linear Technology applications group is available and
extremely qualified to assist in the selection and/or design
of the transformer.
Setting the Current Limit Resistor
The maximum current limit can be set by placing a resistor
between the RILIM pin and ground. This provides some
flexibility in picking standard off-the-shelf transformers that
may be rated for less current than the LT3573’s internal
power switch current limit. If the maximum current limit
is needed, use a 10k resistor. For lower current limits, the
following equation sets the approximate current limit:
R A I k
ILIM LIM
= − +65 10 1 6 10
3
• ( . )
The Switch Current Limit vs RILIM plot in the Typical Per-
formance Characteristics section depicts a more accurate
current limit.
Undervoltage Lockout (UVLO)
The SHDN/UVLO pin is connected to a resistive voltage
divider connected to VIN as shown in Figure 8. The voltage
threshold on the SHDN/UVLO pin for VIN rising is 1.22V.
To introduce hysteresis, the LT3573 draws 2.5µA from the
SHDN/UVLO pin when the pin is below 1.22V. The hysteresis
is therefore user-adjustable and depends on the value of
R1. The UVLO threshold for VIN rising is:
VV R R
RµA R
IN UVLO RISING( , )
. • ( ) . •=++
1 22 1 2
22 5 1
The UVLO threshold for VIN falling is:
VV R R
R
IN UVLO FALLING( , )
. • ( )
=+1 22 1 2
2
To implement external run/stop control, connect a small
NMOS to the UVLO pin, as shown in Figure 8. Turning the
NMOS on grounds the UVLO pin and prevents the LT3573
from operating, and the part will draw less than a 1µA of
quiescent current.
Figure 8. Undervoltage Lockout (UVLO)
LT3573
SHDN/UVLO
GND
R2
R1
VIN
3573 F08
RUN/STOP
CONTROL
(OPTIONAL)
applications inForMation
LT3573
15
3573fc
applications inForMation
values). If too large of an RC value is used, the part will be
more susceptible to high frequency noise and jitter. If too
small of an RC value is used, the transient performance will
suffer. The value choice for CC is somewhat the inverse
of the RC choice: if too small a CC value is used, the loop
may be unstable, and if too large a CC value is used, the
transient performance will also suffer. Transient response
plays an important role for any DC/DC converter.
Design Example
The following example illustrates the converter design
process using LT3573.
Given the input voltage of 20V to 28V, the required output
is 5V, 1A.
VIN(MIN) = 20V, VIN(MAX) = 28V, VOUT = 5V, VF = 0.5V
and IOUT = 1A
1. Select the transformer turns ratio to accommodate
the output.
The output voltage is reflected to the primary side by a
factor of turns ratio N. The switch voltage stress VSW is
expressed as:
NN
N
V V N V V V
P
S
SW MAX IN OUT F
=
= + + <
( ) ( ) 50
Or rearranged to:
NV
V V
IN MAX
OUT F
<−
+
50 ( )
( )
On the other hand, the primary-side current is multiplied by
the same factor of N. The converter output capability is:
I D NI
DN V V
V N
OUT MAX PK
OUT F
IN
( ) . • ( ) •
( )
= −
=+
+
0 8 1 1
2
(( )V V
OUT F
+
Minimum Load Requirement
The LT3573 obtains output voltage information through
the transformer while the secondary winding is conducting
current. During this time, the output voltage (multiplied
times the turns ratio) is presented to the primary side of
the transformer. The LT3573 uses this reflected signal to
regulate the output voltage. This means that the LT3573
must turn on every so often to sample the output voltage,
which delivers a small amount of energy to the output.
This sampling places a minimum load requirement on the
output of 1% to 2% of the maximum load.
BIAS Pin Considerations
For applications with an input voltage less than 15V, the
BIAS pin is typically connected directly to the VIN pin. For
input voltages greater than 15V, it is preferred to leave the
BIAS pin separate form the VIN pin. In this condition, the
BIAS pin is regulated with an internal LDO to a voltage of
3V. By keeping the BIAS pin separate from the input voltage
at high input voltages, the physical size of the capacitors
can be minimized (the BIAS pin can then use a 6.3V or
10V rated capacitor).
Overdriving the BIAS Pin with a Third Winding
The LT3573 provides excellent output voltage regulation
without the need for an opto-coupler, or third winding, but
for some applications with higher input voltages (>20V),
it may be desirable to add an additional winding (often
called a third winding) to improve the system efficiency.
For proper operation of the LT3573, if a winding is used as
a supply for the BIAS pin, ensure that the BIAS pin voltage
is at least 3.15V and always less than the input voltage.
For a typical 24VIN application, overdriving the BIAS pin
will improve the efficiency gain 4-5%.
Loop Compensation
The LT3573 is compensated using an external resistor-
capacitor network on the VC pin. Typical values are in the
range of RC = 50k and CC = 1nF (see the numerous sche-
matics in the Typical Applications section for other possible
LT3573
16
3573fc
applications inForMation
The transformer turns ratio is selected such that the con-
verter has adequate current capability and a switch stress
below 50V. Table 6 shows the switch voltage stress and
output current capability at different transformer turns
ratio.
Table 6. Switch Voltage Stress and Output Current Capability vs
Turns-Ratio
N
VSW(MAX) AT VIN(MAX)
(V)
IOUT(MAX) AT VIN(MIN)
(A)
DUTY CYCLE
(%)
1:1 33.5 0.53 16~22
2:1 39 0.88 28~35
3:1 44.5 1.12 37~45
4:1 50 1.30 44~52
BIAS winding turns ratio is selected to program the BIAS
voltage to 3V~5V. The BIAS voltage shall not exceed the
input voltage.
The turns ratio is then selected as primary: secondary:
BIAS = 3:1:1.
2. Select the transformer primary inductance for target
switching frequency.
The LT3573 requires a minimum amount of time to sample
the output voltage during the off-time. This off-time,
tOFF(MIN), shall be greater than 350ns over all operating
conditions. The converter also has a minimum current limit,
LMIN, of 250mA to help create this off-time. This defines
the minimum required inductance as defined as:
LN V V
It
MIN OUT F
MIN OFF MIN
=+( ) •( )
The transformer primary inductance also affects the
switching frequency which is related to the output ripple. If
above the minimum inductance, the transformer’s primary
inductance may be selected for a target switching frequency
range in order to minimize the output ripple.
The following equation estimates the switching frequency.
ft t I
V
L
I
N V V
L
SW ON OFF PK
IN
PK
PS OUT F
=+=
++
1 1
( )
Table 7.Switching Frequency at Different Primary
Inductance at IPK
L (µH)
fSW AT VIN(MIN)
(kHz)
fSW AT VIN(MAX)
(kHz)
25 236 305
50 121 157
100 61 80
Note: The switching frequency is calculated at maximum output.
In this design example, the minimum primary inductance is
used to achieve a nominal switching frequency of 275kHz
at full load. The PA2454NL from Pulse Engineering is
chosen as the flyback transformer.
Given the turns ratio and primary inductance, a custom-
ized transformer can be designed by magnetic component
manufacturer or a multi-winding transformer such as a
Coiltronics Versa-Pac may be used.
3. Select the output diodes and output capacitor.
The output diode voltage stress VD is the summation of
the output voltage and reflection of input voltage to the
secondary side. The average diode current is the load
current.
V V V
N
D OUT IN
= +
The output capacitor should be chosen to minimize the
output voltage ripple while considering the increase in
size and cost of a larger capacitor. The following equation
calculates the output voltage ripple.
DVLI
CV
MAX PK
OUT
=2
2
4. Select the snubber circuit to clamp the switch
voltage spike.
A flyback converter generates a voltage spike during switch
turn-off due to the leakage inductance of the transformer.
In order to clamp the voltage spike below the maximum
rating of the switch, a snubber circuit is used. There are
many types of snubber circuits, for example R-C, R-C-D and
LT3573
17
3573fc
applications inForMation
Zener clamps. Among them, RCD is widely used. Figure 9
shows the RCD snubber in a flyback converter.
A typical switch node waveform is shown in Figure 10.
During switch turn-off, the energy stored in the leakage
inductance is transferred to the snubber capacitor, and
eventually dissipated in the snubber resistor.
1
2
2
L I f V V N V
R
S PK SW C C OUT
=−( • )
The snubber resistor affects the spike amplitude VC and
duration tSP, the snubber resistor is adjusted such that
tSP is about 150ns. Prolonged tSP may cause distortion
to the output voltage sensing.
The previous steps finish the flyback power stage design.
5. Select the feedback resistor for proper output
voltage.
Using the resistor Tables 1-4, select the feedback resis-
tor RFB, and program the output voltage to 5V. Adjust the
RTC resistor for temperature compensation of the output
voltage. RREF is selected as 6.04k.
A small capacitor in parallel with RREF filters out the noise
during the voltage spike, however, the capacitor should
limit to 10pF. A large capacitor causes distortion on volt-
age sensing.
6. Optimize the compensation network to improve the
transient performance.
The transient performance is optimized by adjusting the
compensation network. For best ripple performance, select
a compensation capacitor not less than 1nF, and select a
compensation resistor not greater than 50k.
7. Current limit resistor, soft-start capacitor and UVLO
resistor divider
Use the current limit resistor RLIM to lower the current
limit if a compact transformer design is required. Soft-start
capacitor helps during the start-up of the flyback converter.
Select the UVLO resistor divider for intended input opera-
tion range. These equations are aforementioned.
Figure 9. RCD Snubber in a Flyback Converter Figure 10. Typical Switch Node Waveform
3573 F09
LS
D
R
+
–C
VIN
VC
NVOUT
tSP 3573 F10
LT3573
18
3573fc
±12V Isolated Flyback Converter
typical applications
SHDN/UVLO
TC
SS
SW
VC GND BIAS
LT3573
3573 TA02
R6
28.7k
R5
10k
VIN
5V VOUT+
5V, 350mA
VOUT–
VIN
3:1 D1
VIN
R1
200k
R2
90.9k
C1
10µFC5
47µF
T1
24µH2.6µH
T1: PULSE PA2454NL OR WÜRTH ELEKTRONIK 750310471
D1: B340A
D2: PMEG6010
C5: MURATA, GRM32ER71A476K
R4
6.04k
R3
80.6k
C2
10nF C3
1000pF
R7
57.6k
R8
2k
D2
C6
0.22µF
TEST
RILIM
RFB
RREF
SHDN/UVLO
TC
SS
SW
VC GND BIAS
LT3573
3573 TA03
R6
59k
R5
10k
VIN
5V
VIN
2:1:1
VIN
R1
200k
R2
90.9k
C1
10µFT1
43.6µH
T1: COILTRONICS VPH1-0076-R
D1, D2: B240A
D3: PMEG6010
C5, C6: MURATA, GRM32ER71A476K
R4
6.04k
R3
118k
VOUT1+
12V, 100mA
VOUT1–
D1
C5
47µF
10.9µH
VOUT2+
VOUT2–
–12V, 100mA
D2
C6
47µF
10.9µH
C2
10nF C3
3300pF
R7
56.2k
R8
2k
D3
C6
0.22µF
TEST
RILIM
RFB
RREF
Low Input Voltage 5V Isolated Flyback Converter
LT3573
19
3573fc
typical applications
VOUT+
5V, 700mA
VOUT–
D1
C5
47µF
2.6µH
T1: PULSE PA2454NL
OR WÜRTH ELEKTRONIK 750370047
D1: B340A
D3: PMEG6010
C5: MURATA, GRM32ER71A476K
SHDN/UVLO
TC
SS
SW
VC GND BIAS
LT3573
3573 TA04
R6
28.7k R5
10k
VIN
12V TO
24V
(*40V) VIN
3:1:1
R1
499k
R2
71.5k
C1
10µFT1
24µH
R4
6.04k
R3
80.6k
C3
1000pF
C4
4.7µF
C2
10nF
R7
45.3k *OPTIONAL THIRD
WINDING FOR
40V OPERATION
D2
L1C
2.6µH
R8
2k
D3
C6
0.22µF
TEST
RILIM
RFB
RREF
5V Isolated Flyback Converter
Efficiency
IOUT (mA)
0
EFFICIENCY (%)
400 800
200 600 1000 1200 1400
3573 TA04b
60
70
80
50
40
10
0
30
90
20
VIN = 12V
VIN = 24V
LT3573
20
3573fc
SHDN/UVLO
TC
SS
SW
VC GND BIAS
LT3573
3573 TA05
R6
19.1k R5
10k
VIN
12V TO 24V
(*40V)
VIN
4:1:1
R1
499k
R2
71.5k
C1
10µFT1
24µH
T1: PULSE PA2362NL
OR COILCRAFT GA3429-BL
D1: B340A
D3: PMEG6010
R4
6.04k
R3
76.8k
VOUT+
3.3V, 1A
VOUT–
D1
C5
47µF
1.5µH
C3
1500pF
C4
4.7µF
C2
10nF
R7
25.5k *OPTIONAL THIRD
WINDING FOR
40V OPERATION
D2
L1C
1.5µH
R8
2k
D3
C6
0.22µF
TEST
RILIM
RFB
RREF
3.3V Isolated Flyback Converter
typical applications
12V Isolated Flyback Converter
SHDN/UVLO
TC
SS SW
VC GND BIAS
LT3573
3573 TA06
R6
59k
R5
10k
VIN 5V VOUT
12V, 400mA
VOUT–
VIN
3:1 D1
VIN
R1
499k
R2
71.5k
C1
10µFC5
47µF
T1
58.5µH6.5µH
T1: COILTRONICS VP1-0102-R
D1: B340A
D2: PMEG6010
R4
6.04k
R3
178k
C2
10nF C3
4700pF
R7
40.2k
R8
2k
D2
C6
0.22µF
TEST
RILIM
RFB
RREF
LT3573
21
3573fc
typical applications
Four Output 12V Isolated Flyback Converter
SHDN/UVLO
TC
SS SW
VC GND BIAS
LT3573
3573 TA07
R6
59k
R5
10k
VIN
12V TO
24V
VIN
2:1:1:1:1
VIN
R1
499k
R2
71.5k
C1
10µFT1
43.6µH
T1: COILTRONICS VPH1-0076-R
D1-D4: B240A
D5: PMEG6010
R4
6.04k
R3
118k
VOUT1+
12V, 60mA
VOUT1–
D1
C5
47µF
10.9µH
VOUT2+
12V, 60mA
VOUT2–
D2
C6
47µF
10.9µH
VOUT3+
12V, 60mA
VOUT3–
D3
C7
47µF
10.9µH
VOUT4+
12V, 60mA
VOUT4–
D4
C8
47µF
10.9µH
C2
10nF C3
0.01µF
R7
20k
R8
2k
D5
C6
0.22µF
TEST
RILIM
RFB
RREF
5V Isolated Flyback Converter Using a Tiny Transformer
SHDN/UVLO
TC
SS SW
VC GND BIAS
LT3573
3573 TA08
R6
28.7k
R5
30k
VIN 12V VOUT
5V, 600mA
VOUT–
VIN
3:1 D1
VIN
R1
200k
R2
90.9k
C1
10µFC5
47µF
T1
20µH2.2µH
T1: BH ELECTRONICS L11-0059
D1: B340A
D2: PMEG6010
R4
6.04k
R3
80.6k
C2
10nF C3
1000pF
R7
47.5k
R8
2k
D2
C6
0.22µF
TEST
RILIM
RFB
RREF
LT3573
22
3573fc
5V Isolated Flyback Converter Using Coupling Inductor
SHDN/UVLO
TC
SS SW
VC GND BIAS
LT3573
3573 TA09
R6
26.1k
R5
10k
VIN
5V VOUT+
5V, 0.2A
VOUT–
VIN
1:1 D1
VIN
R1
200k
R2
90.9k
C1
10µF
C5
47µF
T1
23.6µH23.6µH
T1: BH ELECTRONICS, L10-1022
D1: B220A
D2: CMD5H-3
R4
6.04k
R3
26.1k
C2
10nF C3
1500pF
R7
56.2k
R8
2k
D2
C6
0.22µF
TEST
RILIM
RFB
RREF
typical applications
LT3573
23
3573fc
typical applications
SHDN/UVLO
TC
SS SW
VC GND BIAS
LT3573
3573 TA10
R6
20.5k R5
10k
VIN
6V TO 15V
VOUT+
300V, 5mA
VOUT–
VIN
1:10 D1
VIN
R1
100k
R2
36k
C1
10µF0.056µF
s4
C5, 500V,
T1
100µH
R4
6.04k
R3
150k
C2
10nF
C8
100pF
C3
2200pF
R7
25k
R8
1k
D2
C6
0.22µF
TEST
RILIM
RFB
RREF
1M
T1: WÜRTH ELEKTRONIK 750311681
D1: CMMRIF-06
D2: CMMSHI-60
300V Isolated Flyback Converter
LT3573
24
3573fc
package Description
MSOP (MSE16) 0608 REV A
0.53 p 0.152
(.021 p .006)
SEATING
PLANE
0.18
(.007)
1.10
(.043)
MAX
0.17 –0.27
(.007 – .011)
TYP
0.86
(.034)
REF
0.50
(.0197)
BSC
16
16151413121110
1 2 3 4 5 6 7 8
9
9
18
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
0.254
(.010) 0o – 6o TYP
DETAIL “A”
DETAIL “A”
GAUGE PLANE
5.23
(.206)
MIN
3.20 – 3.45
(.126 – .136)
0.889 p 0.127
(.035 p .005)
RECOMMENDED SOLDER PAD LAYOUT
0.305 p 0.038
(.0120 p .0015)
TYP
0.50
(.0197)
BSC
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 p 0.102
(.112 p .004)
2.845 p 0.102
(.112 p .004)
4.039 p 0.102
(.159 p .004)
(NOTE 3)
1.651 p 0.102
(.065 p .004)
1.651 p 0.102
(.065 p .004)
0.1016 p 0.0508
(.004 p .002)
3.00 p 0.102
(.118 p .004)
(NOTE 4)
0.280 p 0.076
(.011 p .003)
REF
4.90 p 0.152
(.193 p .006)
DETAIL “B”
DETAIL “B”
CORNER TAIL IS PART OF
THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
NO MEASUREMENT PURPOSE
0.12 REF
0.35
REF
MSE Package
16-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1667 Rev A)
LT3573
25
3573fc
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
revision history
REV DATE DESCRIPTION PAGE NUMBER
B 10/09 Replaced Figure 1 10
Updated Typical Applications drawings 18, 21, 22
C 07/10 Added patent numbers and revised Typical Application drawing 1
Revised D1 on Block Diagram 6
Revised Table 5 11
Revised Figure 4 in Applications Information section 13
Revised all drawings in Typical Applications section 18-23
Replaced Related Parts list 26
(Revision history begins at Rev B)
LT3573
26
3573fc
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com
LINEAR TECHNOLOGY CORPORATION 2008
LT 0710 REV C • PRINTED IN USA
relateD parts
typical application
PART NUMBER DESCRIPTION COMMENTS
LT3574 Isolated Flyback Switching Regulator with 60V
Integrated Switch 3V ≤ VIN ≤ 40V, No Opto-Isolator or Third Winding Required,
Up to 3W, MSOP-16
LT3757 Boost, Flyback, SEPIC and Inverting Controller 2.9V ≤ VIN ≤ 40V, Current Mode Control, 100kHz to 1MHz
Programmable Operation Frequency, 3mm × 3mm DFN-10 and
MSOP-10E Package
LT3758 Boost, Flyback, SEPIC and Inverting Controller 5.5V ≤ VIN ≤ 100V, Current Mode Control, 100kHz to 1MHz
Programmable Operation Frequency, 3mm × 3mm DFN-10 and
MSOP-10E Package
LT3837 Isolated No-Opto Synchronous Flyback Controller Ideal for VIN from 4.5V to 36V Limited by External Components,
Up to 60W, Current Mode Control
LT3825 Isolated No-Opto Synchronous Flyback Controller VIN 16V to 75V Limited by External Components, Up to 60W,
Current Mode Control
LT1725 Isolated No-Opto Flyback Controller VIN and VOUT Limited Only by External Components, Ideal for 48V
Nominal Input Voltage
LT1737 Isolated No-Opto Flyback Controller VIN and VOUT Limited Only by External Components, Ideal for 24V
Nominal Input Voltage
LTC
®
1871/LTC1871-1/
LTC1871-7 No RSENSE™ Low Quiescent Current Flyback, Boost and
SEPIC Controller Adjustable Switching Frequency, 2.5V ≤ VIN ≤ 36V,
Burst Mode
®
Operation at Light Loads
LTC3803/LTC3803-3/
LTC3803-5 200kHz/300kHz Flyback DC/DC Controller VIN and VOUT Limited Only by External Components, ThinSOT-6
Package
9V to 30VIN, +5V/–5VOUT Isolated Flyback Converter
SHDN/UVLO
TC
SS SW
VC GND BIAS
LT3573
3573 TA11
R5
28.7k R6
10k
VIN
9V TO 30V
VIN
T1
3:1:1:1
R1
357k
R2
51.1k
C1
10µFL1A
63µH
R4
6.04k
R3
80.6k
VOUT+
+5V, 350mA
COM
VOUT–
–5V, 350mA
D1
C5
47µF
D2
C6
47µF
L1B
7µH
L1C
7µH
C3
2700pF
R7
23.7k
C4
4.7µF
C2
10nF
*OPTIONAL THIRD
WINDING FOR
>24V OPERATION
T1: WÜRTH ELEKTRONIK 750310564
D3
L1D
7µH
R8
2k
D4
C6
0.22µF
TEST
RILIM
RFB
RREF