MIC4125YME TR - MICROCHIP TECHNOLOGY

May 2005 1 M9999-052405

MIC4123/4124/4125 Micrel, Inc.

MIC4123/4124/4125

Dual 3A-Peak Low-Side MOSFET Driver

Bipolar/CMOS/DMOS Process

Features

• Reliable, low-power bipolar/CMOS/DMOS construction

• Latch-up protected to >200mA reverse current

• Logic input withstands swing to –5V

• High 3A-peak output current

• Wide 4.5V to 20V operating range

• Drives 1800pF capacitance in 25ns

• Short <50ns typical delay time

• Delay times consistent with in supply voltage change

• Matched rise and fall times

• TTL logic input independent of supply voltage

• Low equivalent 6pF input capacitance

• Low supply current

3.5mA with logic-1 input

350µA with logic-0 input

• Low 2.3Ω typical output impedance

• Output voltage swings within 25mV of ground or VS.

• ‘426/7/8-, ‘1426/7/8-, ‘4426/7/8-compatible pinout

• Inverting, noninverting, and differential conﬁgurations

• Exposed backside pad packaging reduces heat

ePAD SOIC-8L (θJA = 58°C/W)

4mm x 4mm MLF™-8L (θJA = 45°C/W)

General Description

The MIC4123/4124/4125 family are highly reliable BiC-

MOS/DMOS buffer/driver/MOSFET drivers. They are higher

output current versions of the MIC4126/4127/4128, which

are improved versions of the MIC4426/4427/4428. All three

families are pin-compatible. The MIC4123/4124/4125 driv-

ers are capable of giving reliable service in more demanding

electrical environments than their predecessors. They will

not latch under any conditions within their power and voltage

ratings. They can survive up to 5V of noise spiking, of either

polarity, on the ground pin. They can accept, without either

damage or logic upset, up to half an amp of reverse current

(either polarity) forced back into their outputs.

The MIC4123/4124/4125 series drivers are easier to use, more

ﬂexible in operation, and more forgiving than other CMOS

or bipolar drivers currently available. Their BiCMOS/DMOS

construction dissipates minimum power and provides rail-to-

rail voltage swings.

Primarily intended for driving power MOSFETs, the

MIC4123/4124/4125 drivers are suitable for driving other loads

(capacitive, resistive, or inductive) which require low-imped-

ance, high peak currents, and fast switching times. Heavily

loaded clock lines, coaxial cables, or piezoelectric transducers

are some examples. The only known limitation on loading is

that total power dissipated in the driver must be kept within

the maximum power dissipation limits of the package.

Functional Diagram

Integrated Component Count:

4 Resistors

4 Capacitors

52 Transistors

Ground Unused Inputs

INA

OUTA

INV ER T ING

NONINV ER TIN

0.1mA

0.6mA

2kΩ

INB

OUTB

INV ER T ING

NONINV ER TING

0.1mA

0.6mA

2kΩ

GND

Micrel, Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel + 1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com

MIC4123/4124/4125 Micrel, Inc.

M9999-052405 2 May 2005

Pin & Driver Conﬁguration

Ordering Information

Part Number Package Conﬁguration Lead Finish

MIC4123YME EPAD 8-Lead SOIC Dual Inverting Pb-Free

MIC4123YML 8-Lead MLF Dual Inverting Pb-Free

MIC4124YME EPAD 8-Lead SOIC Dual Non-Inverting Pb-Free

MIC4124YML 8-Lead MLF Dual Non-Inverting Pb-Free

MIC4125YME EPAD 8-Lead SOIC Inverting + Non-Inverting Pb-Free

MIC4125YML 8-Lead MLF Inverting + Non-Inverting Pb-Free

Pin Description

Pin Number Pin Name Function

2 / 4 INA / B Control Input

3 GND Ground: Duplicate pins must be externally connected together

6 VSSupply Input: Duplicate pins must be externally connected together

7 / 5 OUTA / B Output: Duplicate pins must be externally connected together

1, 8 NC Not connected

EP GND Ground: Backside Pad

INA

GND

INB

OUTA

OUTB

MIC4123

Dual

Inverting

MIC4123 MIC4124

INA

GND

INB

OUTA

OUTB

MIC4124

Dual

Noninverting

INA

GND

INB

OUTA

OUTB

MIC4125

Inverting +

Noninverting

MIC4125

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MIC4123/4124/4125 Micrel, Inc.

MIC4123/4124/4125 Electrical Characteristics (Note 4)

4.5V ≤ VS ≤ 20V; TA = 25°C, bold values indicate –40°C ≤ TJ ≤ +125°C; unless noted. Input Voltage slew rate >2.5V/µs.

Symbol Parameter Conditions Min Typ Max Units

Input

VIH Logic 1 Input Voltage 2.4 1.5 V

VIL Logic 0 Input Voltage 1.3 0.8 V

IIN Input Current 0V ≤ VIN ≤ VS –1 1 µA

–10 10 µA

Output

VOH High Output Voltage IOUT = 100µA

VS–0.025

VOL Low Output Voltage IOUT = –100µA 0.025 V

RO Output Resistance HI State IOUT = 10mA, VS = 20V 2.3 5 Ω

8 Ω

Output Resistance LO State IOUT = 10mA, VS = 20V 2.2 5 Ω

8 Ω

IPK Peak Output Current 3 A

I Latch-Up Protection >200 mA

Withstand Reverse Current

Switching Time

tR Rise Time test Figure 1, CL = 1800pF 11 35 ns

60 ns

tF Fall Time test Figure 1, CL = 1800pF 11 35 ns

60 ns

tD1 Delay Tlme test Fﬁgure 1, CL = 1800pF 44 75 ns

100 ns

tD2 Delay Time test Figure 1, CL = 1800pF 59 75 ns

100 ns

Power Supply

IS Power Supply Current VIN = 3.0V (both inputs) 1.5 2.5 mA

3.5 mA

IS Power Supply Current VIN = 0.0V (both inputs) 0.15 0.25 mA

0.3 mA

Note 1. Exceeding the absolute maximum rating may damage the device.

Note 2. The device is not guaranteed to function outside its operating rating.

Note 3. Devices are ESD sensitive. Handling precautions recommended. ESD tested to human body model, 1.5k in series with 100pF.

Note 4. Speciﬁcation for packaged product only.

Absolute Maximum Ratings (Note 1)

Supply Voltage ........................................................... +24V

Input Voltage .................................. VS + 0.3V to GND – 5V

Junction Temperature ................................................150°C

Storage Temperature Range ......................–65°C to 150°C

Lead Temperature (10 sec.) ......................................300°C

ESD Susceptability, Note 3

Operating Ratings (Note 2)

Supply Voltage (VS) ......................................+4.5V to +20V

Junction Temperature Range .....................–40°C to 125°C

Package Thermal Resistance

4mm X 4mm MLF θJA ......................................... 45°C/W

EPAD SOIC-8L θJA ............................................. 58°C/W

MIC4123/4124/4125 Micrel, Inc.

M9999-052405 4 May 2005

Figure 1a. Inverting Driver Switching Time

Test Circuit

INA

INB

MIC4123

OUTA

1800pF

VS = 20V

0.1µF 4.7µF

OUTB

1800pF

INA

INB

MIC4124

OUTA

1800pF

VS = 20V

0.1µF 4.7µF

OUTB

1800pF

tD1

90%

10%

tD2 tR

OU T PUT

INPUT 90%

2.5V

tPW

90%

10%

OU T PUT

INPUT 90%

tD1 tD2

tPW

2.5V

Figure 1b. Non-inverting Driver Switching Time

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MIC4123/4124/4125 Micrel, Inc.

Typical Characteristic Curves

100

120

140

100 1000 10000

RISE TIME (ns)

CAPACITIVE LOAD (pF)

Rise Time

vs. Capactive Load

12V

20V

100

120

140

100 1000 10000

FALL TIME (ns)

CAPACITIVE LOAD (pF)

F a l l T i m e

vs. Capactive Load

12V

20V

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5 10 15 20

OUTPUT RESISTANCE (Ω)

INPUT VOLTAGE (V)

Output Resistance

vs. Supply Voltage

Output Low

Output High

100

5 10 15 20

PROPAGATION DELAY (ns)

SUPPLY VOLTAGE (V)

Propagation Delay

vs. Supply Voltage

td2

td1

CLOAD = 1000pF

5 10 15 20

TIME (ns)

SUPPLY VOLTAGE (V)

Rise and Fall Time

vs. Supply Voltage

R i s e T i m e

F a l l T i me

CLOAD = 1000pF

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MIC4123/4124/4125 Micrel, Inc.

Application Information

Although the MIC4123/24/25 drivers have been speciﬁcally

constructed to operate reliably under any practical circum-

stances, there are nonetheless details of usage which will

provide better operation of the device.

Supply Bypassing

Charging and discharging large capacitive loads quickly

requires large currents. For example, charging 2000pF from

0 to 15 volts in 20ns requires a constant current of 1.5A. In

practice, the charging current is not constant, and will usually

peak at around 3A. In order to charge the capacitor, the driver

must be capable of drawing this much current, this quickly,

from the system power supply. In turn, this means that as far

as the driver is concerned, the system power supply, as seen

by the driver, must have a VERY low impedance.

As a practical matter, this means that the power supply bus

must be capacitively bypassed at the driver with at least

100X the load capacitance in order to achieve optimum

driving speed. It also implies that the bypassing capacitor

must have very low internal inductance and resistance at

all frequencies of interest. Generally, this means using two

capacitors, one a high-performance low ESR ﬁlm, the other

a low internal resistance ceramic, as together the valleys in

their two impedance curves allow adequate performance over

a broad enough band to get the job done. PLEASE NOTE

that many ﬁlm capacitors can be sufﬁciently inductive as to

be useless for this service. Likewise, many multilayer ceramic

capacitors have unacceptably high internal resistance. Use

capacitors intended for high pulse current service (in-house

we use WIMA™ ﬁlm capacitors and AVX Ramguard™ ceram-

ics; several other manufacturers of equivalent devices also

exist). The high pulse current demands of capacitive drivers

also mean that the bypass capacitors must be mounted

very close to the driver in order to prevent the effects of lead

inductance or PCB land inductance from nullifying what you

are trying to accomplish. For optimum results the sum of the

lengths of the leads and the lands from the capacitor body to

the driver body should total 2.5cm or less.

Bypass capacitance, and its close mounting to the driver serves

two purposes. Not only does it allow optimum performance

from the driver, it minimizes the amount of lead length radiat-

ing at high frequency during switching, (due to the large Δ I)

thus minimizing the amount of EMI later available for system

disruption and subsequent cleanup. It should also be noted

that the actual frequency of the EMI produced by a driver is

not the clock frequency at which it is driven, but is related to

the highest rate of change of current produced during switch-

ing, a frequency generally one or two orders of magnitude

higher, and thus more difﬁcult to ﬁlter if you let it permeate your

system. Good bypassing practice is essential to proper

operation of high speed driver ICs.

Grounding

Both proper bypassing and proper grounding are necessary

for optimum driver operation. Bypassing capacitance only

allows a driver to turn the load ON. Eventually (except in rare

circumstances) it is also necessary to turn the load OFF. This

requires attention to the ground path. Two things other than

the driver affect the rate at which it is possible to turn a load

off: The adequacy of the grounding available for the driver,

and the inductance of the leads from the driver to the load.

The latter will be discussed in a separate section.

The E-Pad and MLF packages have an exposed pad under

the package. It's important for good thermal performance that

this pad is connected to a ground plane.

Best practice for a ground path is obviously a well laid out

ground plane. However, this is not always practical, and a

poorly-laid out ground plane can be worse than none. Attention

to the paths taken by return currents even in a ground plane

is essential. In general, the leads from the driver to its load,

the driver to the power supply, and the driver to whatever is

driving it should all be as low in resistance and inductance

as possible. Of the three paths, the ground lead from the

driver to the logic driving it is most sensitive to resistance or

inductance, and ground current from the load are what is most

likely to cause disruption. Thus, these ground paths should

be arranged so that they never share a land, or do so for as

short a distance as is practical.

To illustrate what can happen, consider the following: The

inductance of a 2cm long land, 1.59mm (0.062") wide on a

PCB with no ground plane is approximately 45nH. Assum-

ing a dl/dt of 0.3A/ns (which will allow a current of 3A to ﬂow

after 10ns, and is thus slightly slow for our purposes) a volt-

age of 13.5 Volts will develop along this land in response to

our postulated ∆Ι. For a 1cm land, (approximately 15nH) 4.5

Volts is developed. Either way, anyone using TTL-level input

signals to the driver will ﬁnd that the response of their driver

has been seriously degraded by a common ground path for

input to and output from the driver of the given dimensions.

Note that this is before accounting for any resistive drops in

the circuit. The resistive drop in a 1.59mm (0.062") land of

2oz. Copper carrying 3A will be about 4mV/cm (10mV/in) at

DC, and the resistance will increase with frequency as skin

effect comes into play.

The problem is most obvious in inverting drivers where the

input and output currents are in phase so that any attempt

to raise the driver’s input voltage (in order to turn the driver’s

load off) is countered by the voltage developed on the com-

mon ground path as the driver attempts to do what it was

supposed to. It takes very little common ground path, under

these circumstances, to alter circuit operation drastically.

Output Lead Inductance

The same descriptions just given for PCB land inductance

apply equally well for the output leads from a driver to its load,

except that commonly the load is located much further away

from the driver than the driver’s ground bus.

Generally, the best way to treat the output lead inductance

problem, when distances greater than 4cm (2") are involved,

requires treating the output leads as a transmission line. Un-

fortunately, as both the output impedance of the driver and the

input impedance of the MOSFET gate are at least an order of

magnitude lower than the impedance of common coax, using

coax is seldom a cost-effective solution. A twisted pair works

about as well, is generally lower in cost, and allows use of a

MIC4123/4124/4125 Micrel, Inc.

M9999-052405 7 May 2005

wider variety of connectors. The second wire of the twisted

pair should carry common from as close as possible to the

ground pin of the driver directly to the ground terminal of the

load. Do not use a twisted pair where the second wire in the

pair is the output of the other driver, as this will not provide a

complete current path for either driver. Likewise, do not use

a twisted triad with two outputs and a common return unless

both of the loads to be driver are mounted extremely close

to each other, and you can guarantee that they will never be

switching at the same time.

For output leads on a printed circuit, the general rule is to make

them as short and as wide as possible. The lands should also

be treated as transmission lines: i.e., minimize sharp bends,

or narrowings in the land, as these will cause ringing. For a

rough estimate, on a 1.59mm (0.062") thick G-10 PCB a pair

of opposing lands each 2.36mm (0.093") wide translates to a

characteristic impedance of about 50Ω. Half that width sufﬁces

on a 0.787mm (0.031") thick board. For accurate impedance

matching with a MIC4123/24/25 driver, on a 1.59mm (0.062")

board a land width of 42.75mm (1.683") would be required,

due to the low impedance of the driver and (usually) its load.

This is obviously impractical under most circumstances.

Generally the tradeoff point between lands and wires comes

when lands narrower than 3.18mm (0.125") would be required

on a 1.59mm (0.062") board.

To obtain minimum delay between the driver and the load, it

is considered best to locate the driver as close as possible to

the load (using adequate bypassing). Using matching trans-

formers at both ends of a piece of coax, or several matched

lengths of coax between the driver and the load, works in

theory, but is not optimum.

Driving at Controlled Rates

Occasionally there are situations where a controlled rise or

fall time (which may be considerably longer than the normal

rise or fall time of the driver’s output) is desired for a load. In

such cases it is still prudent to employ best possible practice

in terms of bypassing, grounding and PCB layout, and then

reduce the switching speed of the load (NOT the driver) by

adding a noninductive series resistor of appropriate value

between the output of the driver and the load. For situations

where only rise or only fall should be slowed, the resistor can

be paralleled with a fast diode so that switching in the other

direction remains fast. Due to the Schmitt-trigger action of the

driver’s input it is not possible to slow the rate of rise (or fall)

of the driver’s input signal to achieve slowing of the output.

Input Stage

The input stage of the MIC4123/24/25 consists of a single-

MOSFET class A stage with an input capacitance of <8pF.

This capacitance represents the maximum load from the

driver that will be seen by its controlling logic. The drain load

on the input MOSFET is a –2mA current source. Thus, the

quiescent current drawn by the driver varies, depending upon

the logic state of the input.

Following the input stage is a buffer stage which provides

~400mV of hysteresis for the input, to prevent oscillations

when slowly-changing input signals are used or when noise

is present on the input. Input voltage switching threshold is

approximately 1.5V which makes the driver directly compat-

ible with TTL signals, or with CMOS powered from any supply

voltage between 3V and 15V.

The MIC4123/24/25 drivers can also be driven directly by

the MIC9130, MIC3808, MIC38C42, and similar switch mode

power supply ICs. By relocating the main switch drive function

into the driver rather than using the somewhat limited drive

capabilities of a PWM IC. The PWM IC runs cooler, which

generally improves its performance and longevity, and the

main switches switch faster, which reduces switching losses

and increase system efﬁciency.

The input protection circuitry of the MIC4123/24/25, in addi-

tion to providing 2kV or more of ESD protection, also works to

prevent latchup or logic upset due to ringing or voltage spiking

on the logic input terminal. In most CMOS devices when the

logic input rises above the power supply terminal, or descends

below the ground terminal, the device can be destroyed or

rendered inoperable until the power supply is cycled OFF

and ON. The MIC4123/24/25 drivers have been designed to

prevent this. Input voltages excursions as great as 5V below

ground will not alter the operation of the device. Input excur-

sions above the power supply voltage will result in the excess

voltage being conducted to the power supply terminal of the

IC. Because the excess voltage is simply conducted to the

power terminal, if the input to the driver is left in a high state

when the power supply to the driver is turned off, currents as

high as 30mA can be conducted through the driver from the

input terminal to its power supply terminal. This may overload

the output of whatever is driving the driver, and may cause

other devices that share the driver’s power supply, as well as

the driver, to operate when they are assumed to be off, but

it will not harm the driver itself. Excessive input voltage will

also slow the driver down, and result in much longer internal

propagation delays within the drivers. TD2, for example, may

increase to several hundred nanoseconds. In general, while

the driver will accept this sort of misuse without damage,

proper termination of the line feeding the driver so that line

spiking and ringing are minimized, will always result in faster

and more reliable operation of the device, leave less EMI to

be ﬁltered elsewhere, be less stressful to other components

in the circuit, and leave less chance of unintended modes of

operation.

Power Dissipation

CMOS circuits usually permit the user to ignore power dis-

sipation. Logic families such as 4000 series and 74Cxxx have

outputs which can only source or sink a few milliamps of cur-

rent, and even shorting the output of the device to ground or

VCC may not damage the device. CMOS drivers, on the other

hand, are intended to source or sink several Amps of current.

This is necessary in order to drive large capacitive loads at

frequencies into the megahertz range. Package power dis-

sipation of driver ICs can easily be exceeded when driving

large loads at high frequencies. Care must therefore be paid

to device dissipation when operating in this domain.

May 2005 8 M9999-052405

MIC4123/4124/4125 Micrel, Inc.

The Supply Current vs Frequency and Supply Current vs

Load characteristic curves furnished with this data sheet

aid in estimating power dissipation in the driver. Operating

frequency, power supply voltage, and load all affect power

dissipation.

Given the power dissipation in the device, and the thermal

resistance of the package, junction operating temperature

for any ambient is easy to calculate. For example, the

thermal resistance of the 8-pin E-Pas SOIC package, from

the datasheet, is 58°C/W. In a 25°C ambient, then, using a

maximum junction temperature of 150°C, this package will

dissipate 2.16W.

Accurate power dissipation numbers can be obtained by sum-

ming the three sources of power dissipation in the device:

• Load power dissipation (PL)

• Quiescent power dissipation (PQ)

• Transition power dissipation (PT)

Calculation of load power dissipation differs depending upon

whether the load is capacitive, resistive or inductive.

Resistive Load Power Dissipation

Dissipation caused by a resistive load can be calculated as:

PL = I2 RO D

where:

I = the current drawn by the load

RO = the output resistance of the driver when the

output is high, at the power supply voltage used

(See characteristic curves)

D = fraction of time the load is conducting (duty cycle)

Capacitive Load Power Dissipation

Dissipation caused by a capacitive load is simply the energy

placed in, or removed from, the load capacitance by the

driver. The energy stored in a capacitor is described by the

equation:

E = 1/2 C V2

As this energy is lost in the driver each time the load is charged

or discharged, for power dissipation calculations the 1/2 is

removed. This equation also shows that it is good practice

not to place more voltage in the capacitor than is necessary,

as dissipation increases as the square of the voltage applied

to the capacitor. For a driver with a capacitive load:

PL = f C (VS)2

where:

f = Operating Frequency

C = Load Capacitance

VS = Driver Supply Voltage

Inductive Load Power Dissipation

For inductive loads the situation is more complicated. For

the part of the cycle in which the driver is actively forcing

current into the inductor, the situation is the same as it is in

the resistive case:

PL1 = I2 RO D

However, in this instance the RO required may be either the on

resistance of the driver when its output is in the high state, or

its on resistance when the driver is in the low state, depending

upon how the inductor is connected, and this is still only half

the story. For the part of the cycle when the inductor is forcing

current through the driver, dissipation is best described as

PL2 = I VD (1 – D)

where VD is the forward drop of the clamp diode in the driver

(generally around 0.7V). The two parts of the load dissipation

must be summed in to produce PL

PL = PL1 + PL2

Quiescent Power Dissipation

Quiescent power dissipation (PQ, as described in the input

section) depends on whether the input is high or low. A low

input will result in a maximum current drain (per driver) of

≤0.2mA; a logic high will result in a current drain of ≤2.0mA.

Quiescent power can therefore be found from:

PQ = VS [D IH + (1 – D) IL]

where:

IH = quiescent current with input high

IL = quiescent current with input low

D = fraction of time input is high (duty cycle)

VS = power supply voltage

Transition Power Dissipation

Transition power is dissipated in the driver each time its

output changes state, because during the transition, for a

very brief interval, both the N- and P-channel MOSFETs in

the output totem-pole are ON simultaneously, and a current

is conducted through them from VS to ground. The transition

power dissipation is approximately:

PT = f VS (A•s)

where (A•s) is a time-current factor derived from Figure 2.

Total power (PD) then, as previously described is just

PD = PL + PQ +PT

Examples show the relative magnitude for each term.

EXAMPLE 1: A MIC4123 operating on a 12V supply driving

two capacitive loads of 3000pF each, operating at 250kHz,

with a duty cycle of 50%, in a maximum ambient of 60°C.

First calculate load power loss:

PL = f x C x (VS)2

PL = 250,000 x (3 x 10–9 + 3 x 10–9) x 122

= 0.2160W

Then transition power loss:

PT = f x VS x (A•s)

= 250,000 • 12 • 2.2 x 10–9 = 6.6mW

Then quiescent power loss:

MIC4123/4124/4125 Micrel, Inc.

M9999-052405 9 May 2005

PQ = VS x [D x IH + (1 – D) x IL]

= 12 x [(0.5 x 0.0035) + (0.5 x 0.0003)]

= 0.0228W

Total power dissipation, then, is:

PD = 0.2160 + 0.0066 + 0.0228

= 0.2454W

Assuming an E-Pad SOIC package, with an θJA of 58°C/W,

this will result in the junction running at:

0.2454 x 58 = 14.2°C

above ambient, which, given a maximum ambient tempera-

ture of 60°C, will result in a maximum junction temperature

of 89.4°C.

EXAMPLE 2: A MIC4124 operating on a 15V input, with one

driver driving a 50Ω resistive load at 1MHz, with a duty cycle

of 67%, and the other driver quiescent, in a maximum ambi-

ent temperature of 40°C:

PL = I2 x RO x D

First, IO must be determined.

IO = VS / (RO + RLOAD)

Given RO from the characteristic curves then,

IO = 15 / (3.3 + 50)

IO = 0.281A

and:

PL = (0.281)2 x 3.3 x 0.67

= 0.174W

PT = F x VS x (A•s)/2

(because only one side is operating)

= (1,000,000 x 15 x 3.3 x 10–9) / 2

= 0.025 W

and:

PQ = 15 x [(0.67 x 0.00125) + (0.33 x 0.000125) +

(1 x 0.000125)]

(this assumes that the unused side of the driver has its input

grounded, which is more efﬁcient)

= 0.015W

then:

PD = 0.174 + 0.025 + 0.0150

= 0.213W

In a MLF with an θJA of 60°C/W, this amount of power results

in a junction temperature given the maximum 40°C ambient

of:

(0.213 x 100) + 40 = 52.8°C

The actual junction temperature will be lower than calculated

both because duty cycle is less than 100% and because the

graph lists RDS(on) at a TJ of 125°C and the RDS(on) at 52.8°C TJ

will be somewhat lower.

Deﬁnitions

CL = Load Capacitance in Farads.

D = Duty Cycle expressed as the fraction of time the

input to the driver is high.

f = Operating Frequency of the driver in Hertz.

IH = Power supply current drawn by a driver when both

inputs are high and neither output is loaded.

IL = Power supply current drawn by a driver when both

inputs are low and neither output is loaded.

ID = Output current from a driver in Amps.

PD = Total power dissipated in a driver in Watts.

PL = Power dissipated in the driver due to the driver’s

load

in Watts.

PQ = Power dissipated in a quiescent driver in Watts.

PT = Power dissipated in a driver when the output

changes

states (“shoot-through current”) in Watts.

NOTE: The “shoot-through” current from a dual

transition (once up, once down) for both drivers is

stated in the graph on the following page in ampere-

nanoseconds. This ﬁgure must be multiplied by the

number of repetitions per second (frequency to ﬁnd

Watts).

RO = Output resistance of a driver in Ohms.

VS = Power supply voltage to the IC in Volts.

May 2005 10 M9999-052405

MIC4123/4124/4125 Micrel, Inc.

NOTE: THE VALUES ON THIS GRAPH REPRESENT THE LOSS SEEN BY BOTH

DRIVERS IN A PACKAGE DURING ONE COMPLETE CYCLE. FOR A SINGLE

DRIVER DIVIDE THE STATED VALUES BY 2. FOR A SINGLE TRANSITION OF A

SINGLE DRIVER, DIVIDE THE STATED VALUE BY 4.

Figure 2.

0 2 4 6 8 10 12 14 16 18

A•s (Ampere-seconds)

VIN

C ros s over

E nergy L os s

10-8

10-9

10-10

MIC4123/4124/4125 Micrel, Inc.

M9999-052405 11 May 2005

Package Information

0.20 R.

PIN1 ID

4X P

0PLANE

SEATING

4X P

MAX.

NOM.

MIN.

E2 3.05

b0.23

0.30

0.24

D2 2.25

0.00

3.35

3.20

0.28

0.45

0.65 BSC

0.42

0.35

0.60

12°

0.60

4.00 BSC

3.75 BSC

4.00 BSC

3.75 BSC

2.40

0.85

0.65

0.01

0.20 REF.

2.55

0.80

1.00

0.05

(mm)

DIMENSION

TOP VIEW

BOTTOM VIEW

3. THE PIN#1 IDENTIFIRE MUST BE EXISTED ON THE TOP SURFACE

OF PACKAGE BY USING IDENTIFICATION MARK OR OTHER

FEATURE OF PACKAGE BODY.

4. PACKAGE WARPAGE MAX 0.05mm.

2. N IS THE NUMBER OF TERMINALS.

THE NUMBER OF TERMINALS PER SIDE IS N/4.

NOTE:

1. ALL DIMENSIONS ARE IN MILLIMETERS .

5. APPLIED FOR EXPOSED PAD AND TERMINALS.

8-Pin 4x4 MLF (ML)

8-Pin Exposed Pad SOIC (ME)

MICREL INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA

TEL + 1 (408) 944-0800 FAX + 1 (408) 474-1000 WEB http://www.micrel.com

This information furnished by Micrel in this data sheet is believed to be accurate and reliable. However no responsibility is assumed by Micrel for its use.

Micrel reserves the right to change circuitry and speciﬁcations at any time without notiﬁcation to the customer.

Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can

reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into

the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a signiﬁcant injury to the user. A Purchaser's

use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser's own risk and Purchaser agrees to fully indemnify

Micrel for any damages resulting from such use or sale.