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LM3886 Overture™ Audio Power Amplifier Series
High-Performance 68W Audio Power Amplifier w/Mute
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1FEATURES DESCRIPTION
The LM3886 is a high-performance audio power
23• 68W Cont. Avg. Output Power into 4Ωat VCC =amplifier capable of delivering 68W of continuous
±28V average power to a 4Ωload and 38W into 8Ωwith
• 38W Cont. Avg. Output Power into 8Ωat VCC =0.1% THD+N from 20Hz–20kHz.
±28V The performance of the LM3886, utilizing its Self
• 50W Cont. Avg. Output Power into 8Ωat VCC =Peak Instantaneous Temperature (°Ke) (SPiKe)
±35V protection circuitry, puts it in a class above discrete
• 135W Instantaneous Peak Output Power and hybrid amplifiers by providing an inherently,
Capability dynamically protected Safe Operating Area (SOA).
SPiKe protection means that these parts are
• Signal-to-Noise Ratio ≥92dB completely safeguarded at the output against
• An Input Mute Function overvoltage, undervoltage, overloads, including shorts
• Output Protection from a Short to Ground or to the supplies, thermal runaway, and instantaneous
temperature peaks.
to the Supplies via Internal Current Limiting
Circuitry The LM3886 maintains an excellent signal-to-noise
• Output Over-Voltage Protection against ratio of greater than 92dB with a typical low noise
Transients from Inductive Loads floor of 2.0µV. It exhibits extremely low THD+N
values of 0.03% at the rated output into the rated
• Supply Under-Voltage Protection, not Allowing load over the audio spectrum, and provides excellent
Internal Biasing to Occur when |VEE| + |VCC|≤linearity with an IMD (SMPTE) typical rating of
12V, thus Eliminating Turn-On and Turn-Off 0.004%.
Transients
• 11-Lead TO-220 Package
• Wide Supply Range 20V - 94V
APPLICATIONS
• Component stereo
• Compact stereo
• Self-powered speakers
• Surround-sound amplifiers
• High-end stereo TVs
1Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
2Overture is a trademark of dcl_owner.
3All other trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date. Copyright © 1999–2013, Texas Instruments Incorporated
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
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Typical Application
*Optional components dependent upon specific design requirements. Refer to External Components Description for a
component functional description.
Figure 1. Typical Audio Amplifier Application Circuit
Connection Diagram
Top View
Preliminary: call your local Texas Instruments Sales Rep. or distributor for availability
Figure 2. TO-220 and PFM Plastic Packages
See Package Number NDJ0011B for Staggered Lead Non-Isolated Package or NDA0011B for Staggered
Lead Isolated Package(1)
(1) The LM3886T package TA11B is a non-isolated package, setting the tab of the device and the heat sink at V−potential when the
LM3886 is directly mounted to the heat sink using only thermal compound. If a mica washer is used in addition to thermal compound,
θCS (case to sink) is increased, but the heat sink will be isolated from V−.
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings (1)(2)(3)
Supply Voltage |V+|+|V−| (No Signal) 94V
Supply Voltage |V+|+|V−| (Input Signal) 84V
Common Mode Input Voltage (V+or V−) and
|V+| + |V−|≤80V
Differential Input Voltage (4) 60V
Output Current Internally Limited
Power Dissipation (5) 125W
ESD Susceptibility (6) 3000V
Junction Temperature (7) 150°C
Soldering Information T Package (10 seconds) 260°C
Storage Temperature −40°C to +150°C
Thermal Resistance θJC 1°C/W
θJA 43°C/W
(1) All voltages are measured with respect to the GND pin (pin 7), unless otherwise specified.
(2) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional. Electrical Characteristics state DC and AC electrical specifications under particular test conditions and
specific performance limits. This assumes that the device is within the Operating Ratings. The typical value is a good indication of
device performance.
(3) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
(4) The Differential Input Voltage Absolute Maximum Rating is based on supply voltages of V+= +40V and V−=−40V.
(5) For operating at case temperatures above 25°C, the device must be derated based on a 150°C maximum junction temperature and a
thermal resistance of θJC = 1.0 °C/W (junction to case). Refer to Figure 50 in Application Information under THERMAL
CONSIDERATIONS.
(6) Human body model, 100 pF discharged through a 1.5 kΩresistor.
(7) The operating junction temperature maximum is 150°C, however, the instantaneous Safe Operating Area temperature is 250°C.
Operating Ratings (1)(2)(3)
Temperature Range
TMIN ≤TA≤TMAX −20°C ≤TA≤+85°C
Supply Voltage |V+| + |V−| 20V to 84V
(1) All voltages are measured with respect to the GND pin (pin 7), unless otherwise specified.
(2) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional. Electrical Characteristics state DC and AC electrical specifications under particular test conditions and
specific performance limits. This assumes that the device is within the Operating Ratings. The typical value is a good indication of
device performance.
(3) Operation is specified up to 84V, however, distortion may be introduced from SPIKe Protection Circuitry if proper thermal considerations
are not taken into account. Refer to THERMAL CONSIDERATIONS for more information. (See Figure 5)
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Electrical Characteristics (1)(2)
The following specifications apply for V+= +28V, V−=−28V, IMUTE =−0.5 mA with RL= 4Ωunless otherwise specified. Limits
apply for TA= 25°C. LM3886 Units
Parameter Test Conditions (Limits)
Typical (3) Limit (4)
|V+| + |V−| Power Supply Voltage (5) Vpin7 −V−≥9V 20 V (min)
18 84 V (max)
AMMute Attenuation Pin 8 Open or at 0V, Mute: On
Current out of Pin 8 > 0.5 mA, 115 80 dB (min)
Mute: Off
PO(6) Output Power (Continuous Average) THD + N = 0.1% (max)
f = 1 kHz; f = 20 kHz
|V+| = |V−| = 28V, RL= 4Ω68 60 W (min)
|V+| = |V−| = 28V, RL= 8Ω38 30 W (min)
|V+| = |V−| = 35V, RL= 8Ω50 W
Peak POInstantaneous Peak Output Power 135 W
THD + N Total Harmonic Distortion Plus Noise 60W, RL= 4Ω, %
30W, RL= 8Ω, 0.03 %
20 Hz ≤f≤20 kHz 0.03
AV= 26 dB
SR (6) Slew Rate (7) VIN = 2.0Vp-p, tRISE = 2 ns 19 8 V/μs (min)
I+(6) Total Quiescent Power Supply Current VCM = 0V, Vo= 0V, Io= 0A 50 85 mA (max)
VOS(8) Input Offset Voltage VCM = 0V, Io= 0 mA 1 10 mV (max)
IBInput Bias Current VCM = 0V, Io= 0 mA 0.2 1 μA (max)
IOS Input Offset Current VCM = 0V, Io= 0 mA 0.01 0.2 μA (max)
IoOutput Current Limit |V+| = |V−| = 20V, tON = 10 ms, VO= 0V 11.5 7 A (min)
Vod(8) Output Dropout Voltage (9) |V+–VO|, V+= 28V, Io= +100 mA 1.6 2.0 V (max)
|VO–V−|, V−=−28V, Io=−100 mA 2.5 3.0 V (max)
PSRR (8) Power Supply Rejection Ratio V+= 40V to 20V, V−=−40V, 120 85 dB (min)
VCM = 0V, Io= 0 mA
V+= 40V, V−=−40V to −20V, 105 85 dB (min)
VCM = 0V, Io= 0 mA
CMRR (8) Common Mode Rejection Ratio V+= 60V to 20V, V−=−20V to −60V, 110 85 dB (min)
VCM = 20V to −20V, Io= 0 mA
AVOL(8) Open Loop Voltage Gain |V+| = |V−| = 28V, RL= 2 kΩ,ΔVO= 40V 115 90 dB (min)
GBWP Gain-Bandwidth Product |V+| = |V−| = 30V 8 2 MHz (min)
fO= 100 kHz, VIN = 50 mVrms
eIN(6) Input Noise IHF—A Weighting Filter 2.0 10 μV (max)
RIN = 600Ω(Input Referred)
SNR Signal-to-Noise Ratio PO= 1W, A-Weighted, 92.5 dB
Measured at 1 kHz, RS= 25Ω
PO= 60W, A-Weighted, 110 dB
Measured at 1 kHz, RS= 25Ω
IMD Intermodulation Distortion Test 60 Hz, 7 kHz, 4:1 (SMPTE) 0.004 %
60 Hz, 7 kHz, 1:1 (SMPTE) 0.009
(1) All voltages are measured with respect to the GND pin (pin 7), unless otherwise specified.
(2) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional. Electrical Characteristics state DC and AC electrical specifications under particular test conditions and
specific performance limits. This assumes that the device is within the Operating Ratings. The typical value is a good indication of
device performance.
(3) Typicals are measured at 25°C and represent the parametric norm.
(4) Limits are speficied to AOQL (Average Outgoing Quality Level).
(5) V−must have at least −9V at its pin with reference to ground in order for the under-voltage protection circuitry to be disabled.
(6) AC Electrical Test; refer to Test Circuit #2 -AC Electrical Test Circuit.
(7) The feedback compensation network limits the bandwidth of the closed-loop response and so the slew rate will be reduced due to the
high frequency roll-off. Without feedback compensation, the slew rate is typically larger.
(8) DC Electrical Test; refer to Test Circuit #1- DC Electrical Test Circuit.
(9) The output dropout voltage is the supply voltage minus the clipping voltage. Refer to Figure 14 in Typical Performance Characteristics.
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Single Supply Application Circuit
*Optional components dependent upon specific design requirements. Refer to the External Components Description
for a component functional description.
Figure 3. Typical Single Supply Audio Amplifier Application Circuit
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(excluding active protection circuitry)
External Components Description
(See Figure 1 and Figure 3)
Components Functional Description
1. RIN Acts as a volume control by setting the voltage level allowed to the amplifier's input terminals.
2. RAProvides DC voltage biasing for the single supply operation and bias current for the positive input terminal.
3. CAProvides bias filtering.
4. C Provides AC coupling at the input and output of the amplifier for single supply operation.
5. RBPrevents currents from entering the amplifier's non-inverting input which may be passed through to the load upon power-
down of the system due to the low input impedance of the circuitry when the under-voltage circuitry is off. This
phenomenon occurs when the supply voltages are below 1.5V.
6. CC(1) Reduces the gain (bandwidth of the amplifier) at high frequencies to avoid quasi-saturation oscillations of the output
transistor. The capacitor also suppresses external electromagnetic switching noise created from fluorescent lamps.
7. Ri Inverting input resistance to provide AC Gain in conjunction with Rf1.
8. Ci (1) Feedback capacitor. Ensures unity gain at DC. Also a low frequency pole (highpass roll-off) at:
fc= 1/(2πRi Ci)
9. Rf1 Feedback resistance to provide AC Gain in conjunction with Ri.
10. Rf2(1) At higher frequencies feedback resistance works with Cfto provide lower AC Gain in conjunction with Rf1 and Ri. A high
frequency pole (lowpass roll-off) exists at:
fc= [Rf1 Rf2 (s + 1/Rf2Cf)]/[(Rf1 + Rf2)(s + 1/Cf(Rf1 + Rf2))]
11. Cf(1) Compensation capacitor that works with Rf1 and Rf2 to reduce the AC Gain at higher frequencies.
12. RMMute resistance set up to allow 0.5 mA to be drawn from pin 8 to turn the muting function off.
→RMis calculated using: RM≤(|VEE|−2.6V)/I8 where I8 ≥0.5 mA. Refer to the Figure 44 and Figure 45 in Typical
Performance Characteristics.
13. CMMute capacitance set up to create a large time constant for turn-on and turn-off muting.
14. RSN(1) Works with CSN to stabilize the output stage by creating a pole that eliminates high frequency oscillations.
15. CSN(1) Works with RSN to stabilize the output stage by creating a pole that eliminates high frequency oscillations.
fc= 1/(2πRSNCSN)
16. L (1) Provides high impedance at high frequencies so that R may decouple a highly capacitive load and reduce the Q of the
series resonant circuit due to capacitive load. Also provides a low impedance at low frequencies to short out R and pass
17. R (1) audio signals to the load.
18. CSProvides power supply filtering and bypassing.
19. S1 Mute switch that mutes the music going into the amplifier when opened.
(1) Optional components dependent upon specific design requirements. Refer to Application Information for more information.
OPTIONAL EXTERNAL COMPONENT INTERACTION
Although the optional external components have specific desired functions that are designed to reduce the
bandwidth and eliminate unwanted high frequency oscillations they may cause certain undesirable effects when
they interact. Interaction may occur for components whose reactances are in close proximity to one another. One
example would be the coupling capacitor, CC, and the compensation capacitor, Cf. These two components act as
low impedances to certain frequencies which will couple signals from the input to the output. Please take careful
note of basic amplifier component functionality when designing in these components.
The optional external components shown in Figure 3 and described above are applicable in both single and split
voltage supply configurations.
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Typical Performance Characteristics
SPiKe
Safe Area Protection Response
Figure 4. Figure 5.
Supply Current vs
Supply Voltage Pulse Thermal Resistance
Figure 6. Figure 7.
Supply Current vs
Pulse Thermal Resistance Output Voltage
Figure 8. Figure 9.
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Typical Performance Characteristics (continued)
Pulse Power Limit Pulse Power Limit
Figure 10. Figure 11.
Supply Current vs Input Bias Current vs
Case Temperature Case Temperature
Figure 12. Figure 13.
Clipping Voltage vs Clipping Voltage vs
Supply Voltage Supply Voltage
Figure 14. Figure 15.
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Typical Performance Characteristics (continued)
THD + N THD + N
vs vs
Frequency Frequency
Figure 16. Figure 17.
THD + N THD + N
vs vs
Frequency Output Power
Figure 18. Figure 19.
THD + N THD + N
vs vs
Output Power Output Power
Figure 20. Figure 21.
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Typical Performance Characteristics (continued)
THD + N THD + N
vs vs
Output Power Output Power
Figure 22. Figure 23.
THD + N THD + N
vs vs
Output Power Output Power
Figure 24. Figure 25.
THD + N THD + N
vs vs
Output Power Output Power
Figure 26. Figure 27.
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Typical Performance Characteristics (continued)
THD + N Distribution THD + N Distribution
Figure 28. Figure 29.
THD + N Distribution THD + N Distribution
Figure 30. Figure 31.
Output Power vs
THD + N Distribution Load Resistance
Figure 32. Figure 33.
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Typical Performance Characteristics (continued)
Max Heatsink Thermal Resistance (°C/W)
at the Specified Ambient Temperature (°C)
Maximum Power Dissipation
vs
Supply Voltage
Note: The maximum heat sink thermal resistance values, øSA, in the table above were calculated using a øCS = 0.2°C/W due to
thermal compound. Figure 34.
Power Dissipation Power Dissipation
vs Output Power vs Output Power
Figure 35. Figure 36.
Output Power
vs Supply Voltage IMD 60 Hz, 4:1
Figure 37. Figure 38.
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Typical Performance Characteristics (continued)
IMD 60 Hz, 7 kHz, 4:1 IMD 60 Hz, 7 kHz, 4:1
Figure 39. Figure 40.
IMD 60 Hz, 1:1 IMD 60 Hz, 7 kHz 1:1
Figure 41. Figure 42.
Mute Attenuation vs
IMD 60 Hz, 7 kHz, 1:1 Mute Current
Figure 43. Figure 44.
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Typical Performance Characteristics (continued)
Mute Attenuation vs
Mute Current Large Signal Response
Figure 45. Figure 46.
Power Supply Common-Mode
Rejection Ratio Rejection Ratio
Figure 47. Figure 48.
Open Loop
Frequency Response
Figure 49.
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APPLICATION INFORMATION
GENERAL FEATURES
Mute Function: The muting function of the LM3886 allows the user to mute the music going into the amplifier by
drawing less than 0.5 mA out of pin 8 of the device. This is accomplished as shown in the Typical
Application Circuit where the resistor RMis chosen with reference to your negative supply voltage and is
used in conjuction with a switch. The switch (when opened) cuts off the current flow from pin 8 to V−, thus
placing the LM3886 into mute mode. Refer to Figure 44 and Figure 45 in Typical Performance
Characteristics for values of attenuation per current out of pin 8. The resistance RMis calculated by the
following equation:
RM(|VEE|−2.6V)/I8
where
• I8 ≥0.5 mA. (1)
Under-Voltage Protection: Upon system power-up the under-voltage protection circuitry allows the power
supplies and their corresponding caps to come up close to their full values before turning on the LM3886
such that no DC output spikes occur. Upon turn-off, the output of the LM3886 is brought to ground before
the power supplies such that no transients occur at power-down.
Over-Voltage Protection: The LM3886 contains overvoltage protection circuitry that limits the output current to
approximately 11Apeak while also providing voltage clamping, though not through internal clamping
diodes. The clamping effect is quite the same, however, the output transistors are designed to work
alternately by sinking large current spikes.
SPiKe Protection: The LM3886 is protected from instantaneous peak-temperature stressing by the power
transistor array. Figure 4 in Typical Performance Characteristics shows the area of device operation where
the SPiKe Protection Circuitry is not enabled. The waveform to the right of the SOA graph exemplifies
how the dynamic protection will cause waveform distortion when enabled.
Thermal Protection: The LM3886 has a sophisticated thermal protection scheme to prevent long-term thermal
stress to the device. When the temperature on the die reaches 165°C, the LM3886 shuts down. It starts
operating again when the die temperature drops to about 155°C, but if the temperature again begins to
rise, shutdown will occur again at 165°C. Therefore the device is allowed to heat up to a relatively high
temperature if the fault condition is temporary, but a sustained fault will cause the device to cycle in a
Schmitt Trigger fashion between the thermal shutdown temperature limits of 165°C and 155°C. This
greatly reduces the stress imposed on the IC by thermal cycling, which in turn improves its reliability under
sustained fault conditions.
Since the die temperature is directly dependent upon the heat sink, the heat sink should be chosen as
discussed in THERMAL CONSIDERATIONS, such that thermal shutdown will not be reached during
normal operation. Using the best heat sink possible within the cost and space constraints of the system
will improve the long-term reliability of any power semiconductor device.
THERMAL CONSIDERATIONS
Heat Sinking
The choice of a heat sink for a high-power audio amplifier is made entirely to keep the die temperature at a level
such that the thermal protection circuitry does not operate under normal circumstances. The heat sink should be
chosen to dissipate the maximum IC power for a given supply voltage and rated load.
With high-power pulses of longer duration than 100 ms, the case temperature will heat up drastically without the
use of a heat sink. Therefore the case temperature, as measured at the center of the package bottom, is entirely
dependent on heat sink design and the mounting of the IC to the heat sink. For the design of a heat sink for your
audio amplifier application refer to Determining the Correct Heat Sink.
Since a semiconductor manufacturer has no control over which heat sink is used in a particular amplifier design,
we can only inform the system designer of the parameters and the method needed in the determination of a heat
sink. With this in mind, the system designer must choose his supply voltages, a rated load, a desired output
power level, and know the ambient temperature surrounding the device. These parameters are in addition to
knowing the maximum junction temperature and the thermal resistance of the IC, both of which are provided by
Texas Instruments.
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As a benefit to the system designer we have provided Figure 34 for various loads in Typical Performance
Characteristics, giving an accurate figure for the maximum thermal resistance required for a particular amplifier
design. This data was based on θJC = 1°C/W and θCS = 0.2°C/W. We also provide a section regarding heat sink
determination for any audio amplifier design where θCS may be a different value. It should be noted that the idea
behind dissipating the maximum power within the IC is to provide the device with a low resistance to convection
heat transfer such as a heat sink. Therefore, it is necessary for the system designer to be conservative in his
heat sink calculations. As a rule, the lower the thermal resistance of the heat sink the higher the amount of power
that may be dissipated. This is of course guided by the cost and size requirements of the system. Convection
cooling heat sinks are available commercially, and their manufacturers should be consulted for ratings.
Proper mounting of the IC is required to minimize the thermal drop between the package and the heat sink. The
heat sink must also have enough metal under the package to conduct heat from the center of the package
bottom to the fins without excessive temperature drop.
A thermal grease such as Wakefield type 120 or Thermalloy Thermacote should be used when mounting the
package to the heat sink. Without this compound, thermal resistance will be no better than 0.5°C/W, and
probably much worse. With the compound, thermal resistance will be 0.2°C/W or less, assuming under 0.005
inch combined flatness runout for the package and heat sink. Proper torquing of the mounting bolts is important
and can be determined from heat sink manufacturer's specification sheets.
Should it be necessary to isolate V−from the heat sink, an insulating washer is required. Hard washers like
beryluum oxide, anodized aluminum and mica require the use of thermal compound on both faces. Two-mil mica
washers are most common, giving about 0.4°C/W interface resistance with the compound.
Silicone-rubber washers are also available. A 0.5°C/W thermal resistance is claimed without thermal compound.
Experience has shown that these rubber washers deteriorate and must be replaced should the IC be
dismounted.
Determining Maximum Power Dissipation
Power dissipation within the integrated circuit package is a very important parameter requiring a thorough
understanding if optimum power output is to be obtained. An incorrect maximum power dissipation (PD)
calculation may result in inadequate heat sinking, causing thermal shutdown circuitry to operate and limit the
output power.
The following equations can be used to accurately calculate the maximum and average integrated circuit power
dissipation for your amplifier design, given the supply voltage, rated load, and output power. These equations
can be directly applied to Figure 35 in Typical Performance Characteristics.
Equation 2 exemplifies the maximum power dissipation of the IC and Equation 3,Equation 4 exemplify the
average IC power dissipation expressed in different forms.
PDMAX = VCC2/2π2RL
where
• VCC is the total supply voltage (2)
PDAVE = (VOpk/RL)[VCC/π − VOpk/2]
where
• VCC is the total supply voltage and VOpk = VCC/π(3)
PDAVE = VCC VOpk/πRL−VOpk2/2RL
where
• VCC is the total supply voltage (4)
Determining the Correct Heat Sink
Once the maximum IC power dissipation is known for a given supply voltage, rated load, and the desired rated
output power the maximum thermal resistance (in °C/W) of a heat sink can be calculated. This calculation is
made using Equation 6 and is based on the fact that thermal heat flow parameters are analogous to electrical
current flow properties.
It is also known that typically the thermal resistance, θJC (junction to case), of the LM3886 is 1°C/W and that
using Thermalloy Thermacote thermal compound provides a thermal resistance, θCS (case to heat sink), of about
0.2°C/W as explained in Heat Sinking.
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Referring to the figure below, it is seen that the thermal resistance from the die (junction) to the outside air
(ambient) is a combination of three thermal resistances, two of which are known, θJC and θCS. Since convection
heat flow (power dissipation) is analogous to current flow, thermal resistance is analogous to electrical
resistance, and temperature drops are analogous to voltage drops, the power dissipation out of the LM3886 is
equal to the following:
PDMAX = (TJmax −TAmb)/θJA
where
•θJA =θJC +θCS +θSA (5)
Figure 50.
But since we know PDMAX,θJC, and θSC for the application and we are looking for θSA, we have the following:
θSA = [(TJmax −TAmb)−PDMAX (θJC +θCS)]/PDMAX (6)
Again it must be noted that the value of θSA is dependent upon the system designer's amplifier application and its
corresponding parameters as described previously. If the ambient temperature that the audio amplifier is to be
working under is higher than the normal 25°C, then the thermal resistance for the heat sink, given all other things
are equal, will need to be smaller.
Equation 2 and Equation 6 are the only equations needed in the determination of the maximum heat sink thermal
resistance. This is of course given that the system designer knows the required supply voltages to drive his rated
load at a particular power output level and the parameters provided by the semiconductor manufacturer. These
parameters are the junction to case thermal resistance, θJC, TJmax = 150°C, and the recommended Thermalloy
Thermacote thermal compound resistance, θCS.
SIGNAL-TO-NOISE RATIO
In the measurement of the signal-to-noise ratio, misinterpretations of the numbers actually measured are
common. One amplifier may sound much quieter than another, but due to improper testing techniques, they
appear equal in measurements. This is often the case when comparing integrated circuit designs to discrete
amplifier designs. Discrete transistor amps often “run out of gain” at high frequencies and therefore have small
bandwidths to noise as indicated below.
Integrated circuits have additional open loop gain allowing additional feedback loop gain in order to lower
harmonic distortion and improve frequency response. It is this additional bandwidth that can lead to erroneous
signal-to-noise measurements if not considered during the measurement process. In the typical example above,
the difference in bandwidth appears small on a log scale but the factor of 10 in bandwidth, (200 kHz to 2 MHz)
can result in a 10 dB theoretical difference in the signal-to-noise ratio (white noise is proportional to the square
root of the bandwidth in a system).
In comparing audio amplifiers it is necessary to measure the magnitude of noise in the audible bandwidth by
using a “weighting” filter (see Note below). A “weighting” filter alters the frequency response in order to
compensate for the average human ear's sensitivity to the frequency spectra. The weighting filters at the same
time provide the bandwidth limiting as discussed in the previous paragraph.
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NOTE
CCIR/ARM: A Practical Noise Measurement Method; by Ray Dolby, David Robinson and
Kenneth Gundry, AES Preprint No. 1353 (F-3).
In addition to noise filtering, differing meter types give different noise readings. Meter responses include:
1. RMS reading,
2. average responding,
3. peak reading, and
4. quasi peak reading.
Although theoretical noise analysis is derived using true RMS based calculations, most actual measurements are
taken with ARM (Average Responding Meter) test equipment.
Typical signal-to-noise figures are listed for an A-weighted filter which is commonly used in the measurement of
noise. The shape of all weighting filters is similar, with the peak of the curve usually occurring in the 3 kHz–7 kHz
region as shown below.
Figure 51.
SUPPLY BYPASSING
The LM3886 has excellent power supply rejection and does not require a regulated supply. However, to eliminate
possible oscillations all op amps and power op amps should have their supply leads bypassed with low-
inductance capacitors having short leads and located close to the package terminals. Inadequate power supply
bypassing will manifest itself by a low frequency oscillation known as “motorboating” or by high frequency
instabilities. These instabilities can be eliminated through multiple bypassing utilizing a large tantalum or
electrolytic capacitor (10 μF or larger) which is used to absorb low frequency variations and a small ceramic
capacitor (0.1 μF) to prevent any high frequency feedback through the power supply lines.
If adequate bypassing is not provided the current in the supply leads which is a rectified component of the load
current may be fed back into internal circuitry. This signal causes low distortion at high frequencies requiring that
the supplies be bypassed at the package terminals with an electrolytic capacitor of 470 μF or more.
LEAD INDUCTANCE
Power op amps are sensitive to inductance in the output lead, particularly with heavy capacitive loading.
Feedback to the input should be taken directly from the output terminal, minimizing common inductance with the
load.
Lead inductance can also cause voltage surges on the supplies. With long leads to the power supply, energy is
stored in the lead inductance when the output is shorted. This energy can be dumped back into the supply
bypass capacitors when the short is removed. The magnitude of this transient is reduced by increasing the size
of the bypass capacitor near the IC. With at least a 20 μF local bypass, these voltage surges are important only if
the lead length exceeds a couple feet (>1 μH lead inductance). Twisting together the supply and ground leads
minimizes the effect.
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LAYOUT, GROUND LOOPS AND STABILITY
The LM3886 is designed to be stable when operated at a closed-loop gain of 10 or greater, but as with any other
high-current amplifier, the LM3886 can be made to oscillate under certain conditions. These usually involve
printed circuit board layout or output/input coupling.
When designing a layout, it is important to return the load ground, the output compensation ground, and the low
level (feedback and input) grounds to the circuit board common ground point through separate paths. Otherwise,
large currents flowing along a ground conductor will generate voltages on the conductor which can effectively act
as signals at the input, resulting in high frequency oscillation or excessive distortion. It is advisable to keep the
output compensation components and the 0.1 μF supply decoupling capacitors as close as possible to the
LM3886 to reduce the effects of PCB trace resistance and inductance. For the same reason, the ground return
paths should be as short as possible.
In general, with fast, high-current circuitry, all sorts of problems can arise from improper grounding which again
can be avoided by returning all grounds separately to a common point. Without isolating the ground signals and
returning the grounds to a common point, ground loops may occur.
“Ground Loop” is the term used to describe situations occurring in ground systems where a difference in potential
exists between two ground points. Ideally a ground is a ground, but unfortunately, in order for this to be true,
ground conductors with zero resistance are necessary. Since real world ground leads possess finite resistance,
currents running through them will cause finite voltage drops to exist. If two ground return lines tie into the same
path at different points there will be a voltage drop between them. The first figure below shows a common ground
example where the positive input ground and the load ground are returned to the supply ground point via the
same wire. The addition of the finite wire resistance, R2, results in a voltage difference between the two points as
shown below.
The load current ILwill be much larger than input bias current II, thus V1will follow the output voltage directly, i.e.
in phase. Therefore the voltage appearing at the non-inverting input is effectively positive feedback and the
circuit may oscillate. If there were only one device to worry about then the values of R1and R2would probably be
small enough to be ignored; however, several devices normally comprise a total system. Any ground return of a
separate device, whose output is in phase, can feedback in a similar manner and cause instabilities. Out of
phase ground loops also are troublesome, causing unexpected gain and phase errors.
The solution to most ground loop problems is to always use a single-point ground system, although this is
sometimes impractical. The third figure below is an example of a single-point ground system.
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The single-point ground concept should be applied rigorously to all components and all circuits when possible.
Violations of single-point grounding are most common among printed circuit board designs, since the circuit is
surrounded by large ground areas which invite the temptation to run a device to the closest ground spot. As a
final rule, make all ground returns low resistance and low inductance by using large wire and wide traces.
Occasionally, current in the output leads (which function as antennas) can be coupled through the air to the
amplifier input, resulting in high-frequency oscillation. This normally happens when the source impedance is high
or the input leads are long. The problem can be eliminated by placing a small capacitor, CC, (on the order of 50
pF to 500 pF) across the LM3886 input terminals. Refer to External Components Description relating to
component interaction with Cf.
REACTIVE LOADING
It is hard for most power amplifiers to drive highly capacitive loads very effectively and normally results in
oscillations or ringing on the square wave response. If the output of the LM3886 is connected directly to a
capacitor with no series resistance, the square wave response will exhibit ringing if the capacitance is greater
than about 0.2 μF. If highly capacitive loads are expected due to long speaker cables, a method commonly
employed to protect amplifiers from low impedances at high frequencies is to couple to the load through a 10Ω
resistor in parallel with a 0.7 μH inductor. The inductor-resistor combination as shown in Typical Application
isolates the feedback amplifier from the load by providing high output impedance at high frequencies thus
allowing the 10Ωresistor to decouple the capacitive load and reduce the Q of the series resonant circuit. The LR
combination also provides low output impedance at low frequencies thus shorting out the 10Ωresistor and
allowing the amplifier to drive the series RC load (large capacitive load due to long speaker cables) directly.
GENERALIZED AUDIO POWER AMPLIFIER DESIGN
The system designer usually knows some of the following parameters when starting an audio amplifier design:
Desired Power Output Input Level
Input Impedance Load Impedance
Maximum Supply Voltage Bandwidth
The power output and load impedance determine the power supply requirements, however, depending upon the
application some system designers may be limited to certain maximum supply voltages. If the designer does
have a power supply limitation, he should choose a practical load impedance which would allow the amplifier to
provide the desired output power, keeping in mind the current limiting capabilities of the device. In any case, the
output signal swing and current are found from (where POis the average output power):
(7)
(8)
To determine the maximum supply voltage the following parameters must be considered. Add the dropout
voltage (4V for LM3886) to the peak output swing, Vopeak, to get the supply rail value (i.e. ± (Vopeak + Vod) at a
current of Iopeak). The regulation of the supply determines the unloaded voltage, usually about 15% higher.
Supply voltage will also rise 10% during high line conditions. Therefore, the maximum supply voltage is obtained
from the following equation:
Max. supplies ≊± (Vopeak + Vod)(1 + regulation)(1.1) (9)
The input sensitivity and the output power specs determine the minimum required gain as depicted below:
(10)
Normally the gain is set between 20 and 200; for a 40W, 8Ωaudio amplifier this results in a sensitivity of 894 mV
and 89 mV, respectively. Although higher gain amplifiers provide greater output power and dynamic headroom
capabilities, there are certain shortcomings that go along with the so called “gain.” The input referred noise floor
is increased and hence the SNR is worse. With the increase in gain, there is also a reduction of the power
bandwidth which results in a decrease in feedback thus not allowing the amplifier to respond quickly enough to
nonlinearities. This decreased ability to respond to nonlinearities increases the THD + N specification.
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The desired input impedance is set by RIN. Very high values can cause board layout problems and DC offsets at
the output. The value for the feedback resistance, Rf1, should be chosen to be a relatively large value (10
kΩ–100 kΩ), and the other feedback resistance, Ri, is calculated using standard op amp configuration gain
equations. Most audio amplifiers are designed from the non-inverting amplifier configuration.
DESIGN A 40W/4ΩAUDIO AMPLIFIER
Given:
Power Output 40W
Load Impedance 4Ω
Input Level 1V(max)
Input Impedance 100 kΩ
Bandwidth 20 Hz–20 kHz ± 0.25 dB
Equation 7 and Equation 8 give:
40W/4ΩVopeak = 17.9V Iopeak = 4.5A
Therefore the supply required is: ±21.0V at 4.5A
With 15% regulation and high line the final supply voltage is ±26.6V using Equation 9. At this point it is a good
idea to check the Power Output vs Supply Voltage to ensure that the required output power is obtainable from
the device while maintaining low THD + N. It is also good to check the Power Dissipation vs Supply Voltage to
ensure that the device can handle the internal power dissipation. At the same time designing in a relatively
practical sized heat sink with a low thermal resistance is also important. Refer to Typical Performance
Characteristics graphs and THERMAL CONSIDERATIONS for more information.
The minimum gain from Equation 10 is: AV≥12.6
We select a gain of 13 (Non-Inverting Amplifier); resulting in a sensitivity of 973 mV.
Letting RIN equal 100 kΩgives the required input impedance, however, this would eliminate the “volume control”
unless an additional input impedance was placed in series with the 10 kΩpotentiometer that is depicted in
Figure 1. Adding the additional 100 kΩresistor would ensure the minumum required input impedance.
For low DC offsets at the output we let Rf1 = 100 kΩ. Solving for Ri (Non-Inverting Amplifier) gives the following:
Ri = Rf1/(AV−1) = 100k/(13 −1) = 8.3 kΩ; use 8.2 kΩ
The bandwidth requirement must be stated as a pole, i.e., the 3 dB frequency. Five times away from a pole gives
0.17 dB down, which is better than the required 0.25 dB. Therefore:
fL= 20 Hz/5 = 4 Hz (11)
fH= 20 kHz × 5 = 100 kHz (12)
At this point, it is a good idea to ensure that the Gain-Bandwidth Product for the part will provide the designed
gain out to the upper 3 dB point of 100 kHz. This is why the minimum GBWP of the LM3886 is important.
GBWP ≥AV× f3 dB = 13 × 100 kHz = 1.3 MHz (13)
GBWP = 2.0 MHz (min) for the LM3886 (14)
Solving for the low frequency roll-off capacitor, Ci, we have:
Ci ≥1/(2πRi fL) = 4.85 μF; use 4.7 μF. (15)
Definition of Terms
Input Offset Voltage: The absolute value of the voltage which must be applied between the input terminals
through two equal resistances to obtain zero output voltage and current.
Input Bias Current: The absolute value of the average of the two input currents with the output voltage and
current at zero.
Input Offset Current:The absolute value of the difference in the two input currents with the output voltage and
current at zero.
Input Common-Mode Voltage Range (or Input Voltage Range): The range of voltages on the input terminals
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for which the amplifier is operational. Note that the specifications are not specified over the full common-
mode voltage range unless specifically stated.
Common-Mode Rejection: The ratio of the input common-mode voltage range to the peak-to-peak change in
input offset voltage over this range.
Power Supply Rejection: The ratio of the change in input offset voltage to the change in power supply voltages
producing it.
Quiescent Supply Current: The current required from the power supply to operate the amplifier with no load
and the output voltage and current at zero.
Slew Rate: The internally limited rate of change in output voltage with a large amplitude step function applied to
the input.
Class B Amplifier: The most common type of audio power amplifier that consists of two output devices each of
which conducts for 180° of the input cycle. The LM3886 is a Quasi-AB type amplifier.
Crossover Distortion: Distortion caused in the output stage of a class B amplifier. It can result from inadequate
bias current providing a dead zone where the output does not respond to the input as the input cycle goes
through its zero crossing point. Also for ICs an inadequate frequency response of the output PNP device
can cause a turn-on delay giving crossover distortion on the negative going transition through zero
crossing at the higher audio frequencies.
THD + N: Total Harmonic Distortion plus Noise refers to the measurement technique in which the fundamental
component is removed by a bandreject (notch) filter and all remaining energy is measured including
harmonics and noise.
Signal-to-Noise Ratio: The ratio of a system's output signal level to the system's output noise level obtained in
the absence of a signal. The output reference signal is either specified or measured at a specified
distortion level.
Continuous Average Output Power: The minimum sine wave continuous average power output in watts (or
dBW) that can be delivered into the rated load, over the rated bandwidth, at the rated maximum total
harmonic distortion
Music Power: A measurement of the peak output power capability of an amplifier with either a signal duration
sufficiently short that the amplifier power supply does not sag during the measurement, or when high
quality external power supplies are used. This measurement (an IHF standard) assumes that with normal
music program material the amplifier power supplies will sag insignificantly.
Peak Power: Most commonly referred to as the power output capability of an amplifier that can be delivered to
the load; specified by the part's maximum voltage swing.
Headroom: The margin between an actual signal operating level (usually the power rating of the amplifier with
particular supply voltages, a rated load value, and a rated THD + N figure) and the level just before
clipping distortion occurs, expressed in decibels.
Large Signal Voltage Gain: The ratio of the output voltage swing to the differential input voltage required to
drive the output from zero to either swing limit. The output swing limit is the supply voltage less a specified
quasi-saturation voltage. A pulse of short enough duration to minimize thermal effects is used as a
measurement signal.
Output-Current Limit: The output current with a fixed output voltage and a large input overdrive. The limiting
current drops with time once SPiKe protection circuitry is activated.
Output Saturation Threshold (Clipping Point): The output swing limit for a specified input drive beyond that
required for zero output. It is measured with respect to the supply to which the output is swinging.
Output Resistance: The ratio of the change in output voltage to the change in output current with the output
around zero.
Power Dissipation Rating: The power that can be dissipated for a specified time interval without activating the
protection circuitry. For time intervals in excess of 100 ms, dissipation capability is determined by heat
sinking of the IC package rather than by the IC itself.
Thermal Resistance: The peak, junction-temperature rise, per unit of internal power dissipation (units in °C/W),
above the case temperature as measured at the center of the package bottom.
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The DC thermal resistance applies when one output transistor is operating continuously. The AC thermal
resistance applies with the output transistors conducting alternately at a high enough frequency that the
peak capability of neither transistor is exceeded.
Power Bandwidth: The power bandwidth of an audio amplifier is the frequency range over which the amplifier
voltage gain does not fall below 0.707 of the flat band voltage gain specified for a given load and output
power.
Power bandwidth also can be measured by the frequencies at which a specified level of distortion is
obtained while the amplifier delivers a power output 3 dB below the rated output. For example, an
amplifier rated at 60W with ≤0.25% THD + N, would make its power bandwidth measured as the
difference between the upper and lower frequencies at which 0.25% distortion was obtained while the
amplifier was delivering 30W.
Gain-Bandwidth Product: The Gain-Bandwidth Product is a way of predicting the high-frequency usefulness of
an op amp. The Gain-Bandwidth Product is sometimes called the unity-gain frequency or unity-gain cross
frequency because the open-loop gain characteristic passes through or crosses unity gain at this
frequency. Simply, we have the following relationship: ACL1 × f1= ACL2 × f2
Assuming that at unity-gain (ACL1 = 1 or (0 dB)) fu = fi = GBWP, then we have the following: GBWP = ACL2
× f2
This says that once fu (GBWP) is known for an amplifier, then the open-loop gain can be found at any
frequency. This is also an excellent equation to determine the 3 dB point of a closed-loop gain, assuming
that you know the GBWP of the device. Refer to Figure 52.
Biamplification: The technique of splitting the audio frequency spectrum into two sections and using individual
power amplifiers to drive a separate woofer and tweeter. Crossover frequencies for the amplifiers usually
vary between 500 Hz and 1600 Hz. “Biamping” has the advantages of allowing smaller power amps to
produce a given sound pressure level and reducing distortion effects produced by overdrive in one part of
the frequency spectrum affecting the other part.
C.C.I.R./A.R.M.: Literally: International Radio Consultative Committee/Average Responding Meter
This refers to a weighted noise measurement for a Dolby B type noise reduction system. A filter
characteristic is used that gives a closer correlation of the measurement with the subjective annoyance of
noise to the ear. Measurements made with this filter cannot necessarily be related to unweighted noise
measurements by some fixed conversion factor since the answers obtained will depend on the spectrum
of the noise source.
S.P.L.: Sound Pressure Level—usually measured with a microphone/meter combination calibrated to a pressure
level of 0.0002 μBars (approximately the threshold hearing level).
S.P.L. = 20 Log 10P/0.0002 dB
where P is the R.M.S. sound pressure in microbars. (1 Bar = 1 atmosphere = 14.5 lb/in2= 194 dB S.P.L.).
Figure 52.
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REVISION HISTORY
Changes from Revision B (March 2013) to Revision C Page
• Changed layout of National Data Sheet to TI format .......................................................................................................... 25
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PACKAGE OPTION ADDENDUM
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Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status
(1)
Package Type Package
Drawing Pins Package
Qty Eco Plan
(2)
Lead/Ball Finish
(6)
MSL Peak Temp
(3)
Op Temp (°C) Device Marking
(4/5)
Samples
LM3886T/NOPB ACTIVE TO-220 NDJ 11 20 Green (RoHS
& no Sb/Br) SN Level-1-NA-UNLIM 0 to 70 LM3886T
LM3886TF ACTIVE TO-220 NDA 11 20 TBD Call TI Call TI 0 to 70 LM3886TF
LM3886TF/NOPB ACTIVE TO-220 NDA 11 20 Green (RoHS
& no Sb/Br) SN Level-1-NA-UNLIM 0 to 70 LM3886TF
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
PACKAGE OPTION ADDENDUM
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Addendum-Page 2
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
MECHANICAL DATA
NDJ0011B
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TA11B (Rev B)
MECHANICAL DATA
NDA0011B
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TF11B (Rev D)
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