LM3876 Overture - National Semiconductor

LM3876

Overture

™

Audio Power Amplifier Series

High-Performance 56W Audio Power Amplifier w/Mute

General Description

The LM3876 is a high-performance audio power amplifier

capable of delivering 56W of continuous average power to

an 8Ωload with 0.1%(THD + N) from 20 Hz–20 kHz.

The performance of the LM3876, utilizing its Self Peak In-

stantaneous Temperature (˚Ke) (SPiKe™) Protection Cir-

cuitry, puts it in a class above discrete and hybrid amplifiers

by providing an inherently, dynamically protected Safe Oper-

ating Area (SOA). SPiKe Protection means that these parts

are completely safeguarded at the output against overvolt-

age, undervoltage, overloads, including shorts to the sup-

plies, thermal runaway, and instantaneous temperature

peaks.

The LM3876 maintains an excellent Signal-to-Noise Ratio of

greater than 95 dB(min) with a typical low noise floor of

2.0 µV. It exhibits extremely low (THD + N) values of 0.06%

at the rated output into the rated load over the audio spec-

trum, and provides excellent linearity with an IMD (SMPTE)

typical rating of 0.004%.

Features

n56W continuous average output power into 8Ω

n100W instantaneous peak output power capability

nSignal-to-Noise Ratio ≥95 dB(min)

nAn input mute function

nOutput protection from a short to ground or to the

supplies via internal current limiting circuitry

nOutput over-voltage protection against transients from

inductive loads

nSupply under-voltage protection, not allowing internal

biasing to occur when |V

|+|V

|≤12V, thus

eliminating turn-on and turn-off transients

n11-lead TO-220 package

Applications

nComponent stereo

nCompact stereo

nSelf-powered speakers

nSurround-sound amplifiers

nHigh-end stereo TVs

Overture™and SPiKe™Protection are trademarks of National Semiconductor Corporation.

April 1995

LM3876 Overture Audio Power Amplifier Series

High-Performance 56W Audio Power Amplifier w/Mute

Typical Application

Connection Diagram

DS011832-1

*Optional components dependent upon specific design requirements. Refer to the External Components Description section for a component functional

description.

FIGURE 1. Typical Audio Amplifier Application Circuit

Plastic Package (Note 11)

DS011832-2

†Connect Pin 5 to V+for Compatibility with LM3886.

Top View

Order Number LM3876T

or LM3876TF

See NS Package Number TA11B for

Staggered Lead Non-Isolated

Package or TF11B for

Staggered Lead Isolated Package

www.national.com 2

Absolute Maximum Ratings (Notes 4, 5)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

Supply Voltage |V

|+|V

−

| (No Signal) 94V

Supply Voltage |V

|+|V

−

| (Input Signal) 84V

Common Mode Input Voltage (V

or V

−

) and

|+|V

−

|≤80V

Differential Input Voltage 60V

Output Current Internally Limited

Power Dissipation (Note 6) 125W

ESD Susceptibility (Note 7) 3000V

Junction Temperature (Note 8) 150˚C

Soldering Information

T Package (10 seconds) 260˚C

Storage Temperature −40˚C to +150˚C

Thermal Resistance

1˚C/W

43˚C/W

Operating Ratings (Notes 4, 5)

Temperature Range

MIN

≤T

MAX

−20˚C ≤T

≤+85˚C

Supply Voltage |V

|+|V

−

| 24V to 84V

Note 1: Operation is guaranteed up to 84V, however, distortion may be intro-

duced from SPiKe Protection Circuitry when operating above 70V if proper

thermal considerations are not taken into account. Refer to the Thermal

Considerations section for more information.

(See SPiKe Protection Response)

Electrical Characteristics (Notes 4, 5)

The following specifications apply for V

=+35V, V

−

=−35V, I

MUTE

=−0.5 mA with R

=8Ωunless otherwise specified. Limits

apply for T

=25˚C.

Symbol Parameter Conditions LM3876 Units

(Limits)

Typical Limit

(Note 9) (Note 10)

|+|V

−

| Power Supply Voltage (Note 13) V

pin7

−V

−

≥9V 18 24 V (min)

84 V (max)

Mute Attenuation Pin 8 Open or at 0V, Mute: On

Current out of Pin 8 >0.5 mA, 120 80 dB (min)

Mute: Off

(Note 3) Output Power (Continuous Average) THD + N =0.1%(max) 56 40 W (min)

f=1 kHz; f =20 kHz

Peak P

Instantaneous Peak Output Power 100 W

THD + N Total Harmonic Distortion Plus Noise 40W, 20 Hz ≤f≤20 kHz 0.06 %

=26 dB

(Note 3) Slew Rate (Note 12) V

=1.2 Vrms, f =10 kHz, 11 5 V/µs

(min)

Square-Wave, R

=2kΩ

(Note 2) Total Quiescent Power Supply

Current V

=0V, V

=0V, I

=0A, I

mute

=0A 30 70 mA (max)

(Note 2) Input Offset Voltage V

=0V, I

=0 mA 1 15 mV (max)

Input Bias Current V

=0V, I

=0 mA 0.2 1 µA (max)

Input Offset Current V

=0V, I

=0 mA 0.01 0.2 µA (max)

Output Current Limit |V

|=|V

−

|=12V, t

=10 ms, V

=0V 6 4 A (min)

(Note 2) Output Dropout Voltage (Note 14) |V

–V

|, V

=20V, I

=+100 mA 1.6 5 V (max)

–V

−

|, V

−

=−20V, I

=−100 mA 2.7 5 V (max)

PSRR

(Note 2) Power Supply Rejection Ratio V

=40V to 20V, V

−

=−40V, 120 85 dB (min)

=0V, I

=0mA

=40V, V

−

=−40V to −20V, 120 85 dB (min)

=0V, I

=0mA

CMRR

(Note 2) Common Mode Rejection Ratio V

=60V to 20V, V

−

=−20V to −60V, 120 80 dB (min)

=20V to −20V, I

=0mA

VOL

(Note 2) Open Loop Voltage Gain |V

|=|V

−

|=40V, R

=2kΩ,∆V

=60V 120 90 dB (min)

GBWP Gain-Bandwidth Product |V

|=|V

−

|=40V 8 2 MHz

(min)

=100 kHz, V

=50 mVrms

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Electrical Characteristics (Notes 4, 5) (Continued)

The following specifications apply for V

=+35V, V

−

=−35V, I

MUTE

=−0.5 mA with R

=8Ωunless otherwise specified. Limits

apply for T

=25˚C.

Symbol Parameter Conditions LM3876 Units

(Limits)

Typical Limit

(Note 9) (Note 10)

(Note 3) Input Noise IHF—A Weighting Filter 2.0 8 µV (max)

=600Ω(Input Referred)

SNR Signal-to-Noise Ratio P

=1W, A-Weighted, 98 dB

Measured at 1 kHz, R

=25Ω

=40W, A-Weighted, 114 dB

Measured at 1 kHz, R

=25Ω

Ppk=100W, A-Weighted, 122 dB

Measured at 1 kHz, R

=25Ω

IMD Intermodulation Distortion Test 60 Hz, 7 kHz, 4:1 (SMPTE) 0.004 %

60 Hz, 7 kHz, 1:1 (SMPTE) 0.006

Note 2: DC Electrical Test; refer to Test Circuit #1.

Note 3: AC Electrical Test; refer to Test Circuit #2.

Note 4: All voltages are measured with respect to the GND pin (pin 7), unless otherwise specified.

Note 5: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is

functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions which

guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters where no limit

is given, however, the typical value is a good indication of device performance.

Note 6: For operating at case temperatures above 25˚C, the device must be derated based on a 150˚C maximum junction temperature and a thermal resistance of

θJC =1.0 ˚C/W (junction to case). Refer to the Thermal Resistance figure in the Application Information section under Thermal Considerations.

Note 7: Human body model, 100 pF discharged through a 1.5 kΩresistor.

Note 8: The operating junction temperature maximum is 150˚C, however, the instantaneous Safe Operating Area temperature is 250˚C.

Note 9: Typicals are measured at 25˚C and represent the parametric norm.

Note 10: Limits are guaranteed to National’s AOQL (Average Outgoing Quality Level).

Note 11: The LM3876T package TA11B is a non-isolated package, setting the tab of the device and the heat sink at V−potential when the LM3876 is directly

mounted to the heat sink using only thermal compound. If a mica washer is used in addition to thermal compound, θCS (case to sink) is increased, but the heat sink

will be isolated from V−.

Note 12: The feedback compensation network limits the bandwidth of the closed-loop response and so the slew rate will be reduced due to the high frequency

roll-off. Without feedback compensation, the slew rate is typically 16V/µs.

Note 13: V−must have at least −9V at its pin with reference to ground in order for the under-voltage protection circuitry to be disabled.

Note 14: The output dropout voltage is the supply voltage minus the clipping voltage. Refer to the Clipping Voltage vs Supply Voltage graph in the Typical Perfor-

mance Characteristics section.

Test Circuit #1(DC Electrical Test Circuit)

DS011832-3

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Test Circuit #2(AC Electrical Test Circuit)

Single Supply Application Circuit

DS011832-4

DS011832-5

*Optional components dependent upon specific design requirements. Refer to the External

Components Description section for a component functional description.

FIGURE 2. Typical Single Supply Audio Amplifier Application Circuit

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Equivalent Schematic (excluding active protection circuitry)

External Components Description

(

Figures 1, 2

)

Components Functional Description

1. R

Acts as a volume control by setting the voltage level allowed to the amplifier’s input terminals.

2. R

Provides DC voltage biasing for the single supply operation and bias current for the positive input terminal.

3. C

Provides bias filtering.

4. C Provides AC coupling at the input and output of the amplifier for single supply operation.

5. R

Prevents currents from entering the amplifier’s non-inverting input which may be passed through to the load

upon power-down of the system due to the low input impedance of the circuitry when the under-voltage

circuitry is off. This phenomenon occurs when the supply voltages are below 1.5V.

6. C

(Note 15) Reduces the gain (bandwidth of the amplifier) at high frequencies to avoid quasi-saturation oscillations of

the output transistor. The capacitor also suppresses external electromagnetic switching noise created from

fluorescent lamps.

7. Ri Inverting input resistance to provide AC Gain in conjunction with R

8. Ci

(Note 15) Feedback capacitor. Ensures unity gain at DC. Also a low frequency pole (highpass roll-off) at:

=1/(2πRi Ci)

9. R

Feedback resistance to provide AC Gain in conjunction with Ri.

DS011832-6

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External Components Description (Continued)

(

Figures 1, 2

)

Components Functional Description

10. R

(Note 15) At higher frequencies feedback resistance works with C

to provide lower AC Gain in conjunction with R

and Ri. A high frequency pole (lowpass roll-off) exists at:

=[R

(s + 1/R

)]/[(R

)(s + 1/C

))]

11. C

(Note 15) Compensation capacitor that works with R

and R

to reduce the AC Gain at higher frequencies.

12. R

Mute resistance set up to allow 0.5 mA to be drawn from pin 8 to turn the muting function off.

→R

is calculated using: R

≤(|V

| − 2.6V)/I8 where I8 ≥0.5 mA. Refer to the Mute Attenuation vs

Mute Current curves in the Typical Performance Characteristics section.

13. C

Mute capacitance set up to create a large time constant for turn-on and turn-off muting.

14. R

(Note 15) Works with C

to stabilize the output stage by creating a pole that eliminates high frequency oscillations.

15. C

(Note 15) Works with R

to stabilize the output stage by creating a pole that eliminates high frequency oscillations.

=1/(2πR

)

16. L

(Note 15) Provides high impedance at high frequecies so that R may decouple a highly capacitive load

17. R

(Note 15) and reduce the Q of the series resonant circuit due to capacitive load. Also provides a low impedance at

low frequencies to short out R and pass audio signals to the load.

18. C

Provides power supply filtering and bypassing.

19. S1 Mute switch that mutes the music going into the amplifier when opened.

Note 15: Optional components dependent upon specific design requirements. Refer to the Application Information section for more information.

OPTIONAL EXTERNAL COMPONENT INTERACTION

Although the optional external components have specific desired functions that are designed to reduce the bandwidth and elimi-

nate unwanted high frequency oscillations they may cause certain undesirable effects when they interact. Interaction may occur

for components whose reactances are in close proximity to one another. One example would be the coupling capacitor, C

, and

the compensation capacitor, Cf. These two components act as low impedances to certain frequencies which will couple signals

from the input to the output. Please take careful note of basic amplifier component functionality when designing in these compo-

nents.

The optional external components shown in

Figure 2

and described above are applicable in both single and split voltage supply

configurations.

Typical Performance Characteristics

Safe Area

DS011832-17

SPiKe

Protection Response

DS011832-18

Supply Current vs

Supply Voltage

DS011832-19

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Typical Performance Characteristics (Continued)

Pulse Thermal

Resistance

DS011832-20

Pulse Thermal

Resistance

DS011832-21

Supply Current vs

Output Voltage

DS011832-22

Pulse Power Limit

DS011832-23

Pulse Power Limit

DS011832-24

Supply Current vs

Case Temperature

DS011832-25

Pulse Response

DS011832-26

Input Bias Current vs

Case Temperature

DS011832-27

Clipping Voltage vs

Supply Voltage

DS011832-28

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Typical Performance Characteristics (Continued)

THD+N vs Frequency

DS011832-29

THD+N vs Output Power

DS011832-30

THD+N vs Output Power

DS011832-31

THD+N Distribution

DS011832-32

THD+N Distribution

DS011832-33

Output Power vs

Load Resistance

DS011832-34

Max Heatsink Thermal Resistance (˚C/W)

at the Specified Ambient Temperature (˚C)

PDmaxrvs Supply Voltage

DS011832-9

Note: The maximum heat sink thermal resistance values, øSA, in the table above were calculated using a øCS = 0.2˚C/W due to thermal compound

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Typical Performance Characteristics (Continued)

Power Dissipation vs

Output Power

DS011832-35

Power Dissipation vs

Output Power

DS011832-36

Output Power vs

Supply Voltage

DS011832-37

IMD 60 Hz, 4:1

DS011832-38

IMD 60 Hz, 7 kHz, 4:1

DS011832-39

IMD 60 Hz, 7 kHz, 4:1

DS011832-40

IMD 60 Hz, 1:1

DS011832-41

IMD 60 Hz, 7 kHz 1:1

DS011832-42

IMD 60 Hz, 7 kHz, 1:1

DS011832-43

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Typical Performance Characteristics (Continued)

Application Information

GENERAL FEATURES

Mute Function: The muting function of the LM3876 allows

the user to mute the music going into the amplifier by draw-

ing less than 0.5 mAout of pin 8 of the device. This is accom-

plished as shown in the Typical Application Circuit where the

resistor R

is chosen with reference to your negative supply

voltage and is used in conjuction with a switch. The switch

(when opened) cuts off the current flow from pin 8 to V

−

, thus

placing the LM3876 into mute mode. Refer to the Mute At-

tenuation vs Mute Current curves in the Typical Perfor-

mance Characteristics section for values of attenuation per

current out of pin 8. The resistance R

is calculated by the

following equation:

(|V

| − 2.6V)/I8 where I8 ≥0.5 mA.

Under-Voltage Protection: Upon system power-up the

under-voltage protection circuitry allows the power supplies

and their corresponding caps to come up close to their full

values before turning on the LM3876 such that no DC output

spikes occur. Upon turn-off, the output of the LM3876 is

brought to ground before the power supplies such that no

transients occur at power-down.

Over-Voltage Protection: The LM3876 contains overvolt-

age protection circuitry that limits the output current to ap-

proximately 6Apeak while also providing voltage clamping,

though not through internal clamping diodes. The clamping

effect is quite the same, however, the output transistors are

designed to work alternately by sinking large current spikes.

SPiKe Protection: The LM3876 is protected from instanta-

neous peak-temperature stressing by the power transistor

array. The Safe OperatingArea graph in the Typical Perfor-

mance Characteristics section shows the area of device

operation where the SPiKe Protection Circuitry is not en-

abled. The waveform to the right of the SOA graph exempli-

fies how the dynamic protection will cause waveform distor-

tion when enabled.

Thermal Protection: The LM3876 has a sophisticated ther-

mal protection scheme to prevent long-term thermal stress

to the device. When the temperature on the die reaches

165˚C, the LM3876 shuts down. It starts operating again

when the die temperature drops to about 155˚C, but if the

temperature again begins to rise, shutdown will occur again

at 165˚C. Therefore the device is allowed to heat up to a

relatively high temperature if the fault condition is temporary,

but a sustained fault will cause the device to cycle in a

Schmitt Trigger fashion between the thermal shutdown tem-

perature limits of 165˚C and 155˚C. This greatly reduces the

stress imposed on the IC by thermal cycling, which in turn

improves its reliability under sustained fault conditions.

Since the die temperature is directly dependent upon the

heat sink, the heat sink should be chosen as discussed in

the Thermal Considerations section, such that thermal

shutdown will not be reached during normal operation. Using

the best heat sink possible within the cost and space con-

straints of the system will improve the long-term reliability of

any power semiconductor device.

Mute Attenuation vs

Mute Current

DS011832-44

Mute Attenuation vs

Mute Current

DS011832-45

Large Signal Response

DS011832-46

Power Supply

Rejection Ratio

DS011832-47

Common-Mode

Rejection Ratio

DS011832-48

Open Loop

Frequency Response

DS011832-49

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Application Information (Continued)

THERMAL CONSIDERATIONS

Heat Sinking

The choice of a heat sink for a high-power audio amplifier is

made entirely to keep the die temperature at a level such

that the thermal protection circuitry does not operate under

normal circumstances. The heat sink should be chosen to

dissipate the maximum IC power for a given supply voltage

and rated load.

With high-power pulses of longer duration than 100 ms, the

case temperature will heat up drastically without the use of a

heat sink. Therefore the case temperature, as measured at

the center of the package bottom, is entirely dependent on

heat sink design and the mounting of the IC to the heat sink.

For the design of a heat sink for your audio amplifier applica-

tion refer to the Determining The Correct Heat Sink sec-

tion.

Since a semiconductor manufacturer has no control over

which heat sink is used in a particular amplifier design, we

can only inform the system designer of the parameters and

the method needed in the determination of a heat sink. With

this in mind, the system designer must choose his supply

voltages, a rated load, a desired output power level, and

know the ambient temperature surrounding the device.

These parameters are in addition to knowing the maximum

junction temperature and the thermal resistance of the IC,

both of which are provided by National Semiconductor.

As a benefit to the system designer we have provided Maxi-

mum Power Dissipation vs Supply Voltages curves for vari-

ous loads in the Typical Performance Characteristics sec-

tion, giving an accurate figure for the maximum thermal

resistance required for a particular amplifier design. This

data was based on θ

=1˚C/W and θ

=0.2˚C/W. We also

provide a section regarding heat sink determination for any

audio amplifier design where θ

may be a different value. It

should be noted that the idea behind dissipating the maxi-

mum power within the IC is to provide the device with a low

resistance to convection heat transfer such as a heat sink.

Therefore, it is necessary for the system designer to be con-

servative in his heat sink calculations. As a rule, the lower

the thermal resistance of the heat sink the higher the amount

of power that may be dissipated. This is of course guided by

the cost and size requirements of the system. Convection

cooling heat sinks are available commercially, and their

manufacturers should be consulted for ratings.

Proper mounting of the IC is required to minimize the thermal

drop between the package and the heat sink. The heat sink

must also have enough metal under the package to conduct

heat from the center of the package bottom to the fins with-

out excessive temperature drop.

A thermal grease such as Wakefield type 120 or Thermalloy

Thermacote should be used when mounting the package to

the heat sink. Without this compound, thermal resistance will

be no better than 0.5˚C/W, and probably much worse. With

the compound, thermal resistance will be 0.2˚C/W or less,

assuming under 0.005 inch combined flatness runout for the

package and heat sink. Proper torquing of the mounting

bolts is important and can be determined from heat sink

manufacturer’s specification sheets.

Should it be necessary to isolate V

−

from the heat sink, an in-

sulating washer is required. Hard washers like beryluum ox-

ide, anodized aluminum and mica require the use of thermal

compound on both faces. Two-mil mica washers are most

common, giving about 0.4˚C/W interface resistance with the

compound.

Silicone-rubber washers are also available. A 0.5˚C/W ther-

mal resistance is claimed without thermal compound. Expe-

rience has shown that these rubber washers deteriorate and

must be replaced should the IC be dismounted.

Determining Maximum Power Dissipation

Power dissipation within the integrated circuit package is a

very important parameter requiring a thorough understand-

ing if optimum power output is to be obtained. An incorrect

maximum power dissipation (P

) calculation may result in in-

adequate heat sinking, causing thermal shutdown circuitry to

operate and limit the output power.

The following equations can be used to acccurately calculate

the maximum and average integrated circuit power dissipa-

tion for your amplifier design, given the supply voltage, rated

load, and output power. These equations can be directly ap-

plied to the Power Dissipation vs Output Power curves in the

Typical Performance Characteristics section.

Equation (1)

exemplifies the maximum power dissipation of

the IC and

Equations (2), (3)

exemplify the average IC power

dissipation expressed in different forms.

DMAX

2/2π

(1)

where V

is the total supply voltage

DAVE

=(V

Opk

)[V

/π−V

Opk

/2] (2)

where V

is the total supply voltage and V

Opk

/π

DAVE

Opk

/πR

−V

Opk2

/2R

(3)

where V

is the total supply voltage.

Determining the Correct Heat Sink

Once the maximum IC power dissipation is known for a

given supply voltage, rated load, and the desired rated out-

put power the maximum thermal resistance (in ˚C/W) of a

heat sink can be calculated. This calculation is made using

Equation (4)

and is based on the fact that thermal heat flow

parameters are analogous to electrical current flow proper-

ties.

It is also known that typically the thermal resistance, θ

(junction to case), of the LM3876 is 1˚C/W and that using

Thermalloy Thermacote thermal compound provides a ther-

mal resistance, θ

(case to heat sink), of about 0.2˚C/W as

explained in the Heat Sinking section.

Referring to the figure below, it is seen that the thermal resis-

tance from the die (junction) to the outside air (ambient) is a

combination of three thermal resistances, two of which are

known, θ

and θ

. Since convection heat flow (power dis-

sipation) is analogous to current flow, thermal resistance is

analogous to electrical resistance, and temperature drops

are analogous to voltage drops, the power dissipation out of

the LM3876 is equal to the following:

DMAX

=(T

Jmax

−T

Amb

)/θ

where θ

=θ

+θ

DS011832-12

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Application Information (Continued)

But since we know P

DMAX

,θ

, and θ

for the application

and we are looking for θ

, we have the following:

=[(T

Jmax

−T

Amb

)−P

DMAX

(θ

+θ

)]/P

DMAX

(4)

Again it must be noted that the value of θ

is dependent

upon the system designer’s amplifier application and its cor-

responding parameters as described previously. If the ambi-

ent temperature that the audio amplifier is to be working un-

der is higher than the normal 25˚C, then the thermal

resistance for the heat sink, given all other things are equal,

will need to be smaller.

Equations (1), (4)

are the only equations needed in the de-

termination of the maximum heat sink thermal resistance.

This is of course given that the system designer knows the

required supply voltages to drive his rated load at a particular

power output level and the parameters provided by the semi-

conductor manufacturer. These parameters are the junction

to case thermal resistance, θ

Jmax

=150˚C, and the rec-

ommended Thermalloy Thermacote thermal compound re-

sistance, θ

SIGNAL-TO-NOISE RATIO

In the measurement of the signal-to-noise ratio, misinterpre-

tations of the numbers actually measured are common. One

amplifier may sound much quieter than another, but due to

improper testing techniques, they appear equal in measure-

ments. This is often the case when comparing integrated cir-

cuit designs to discrete amplifier designs. Discrete transistor

amps often “run out of gain” at high frequencies and there-

fore have small bandwidths to noise as indicated below.

Integrated circuits have additional open loop gain allowing

additional feedback loop gain in order to lower harmonic dis-

tortion and improve frequency response. It is this additional

bandwidth that can lead to erroneous signal-to-noise mea-

surements if not considered during the measurement pro-

cess. In the typical example above, the difference in band-

width appears small on a log scale but the factor of 10 in

bandwidth, (200 kHz to 2 MHz) can result in a 10 dB theoreti-

cal difference in the signal-to-noise ratio (white noise is pro-

portional to the square root of the bandwidth in a system).

In comparing audio amplifiers it is necessary to measure the

magnitude of noise in the audible bandwidth by using a

“weighting” filter (Note 16). A “weighting” filter alters the fre-

quency response in order to compensate for the average hu-

man ear’s sensitivity to the frequency spectra. The weighting

filters at the same time provide the bandwidth limiting as dis-

cussed in the previous paragraph.

Note 16: CCIR/ARM:

A Practical Noise Measurement Method;

by Ray

Dolby, David Robinson and Kenneth Gundry, AES Preprint No. 1353 (F-3).

In addition to noise filtering, differing meter types give differ-

ent noise readings. Meter responses include:

1. RMS reading,

2. average responding,

3. peak reading, and

4. quasi peak reading.

Although theoretical noise analysis is derived using true

RMS based calculations, most actual measurements are

taken with ARM (Average Responding Meter) test equip-

ment.

Typical signal-to-noise figures are listed for anA-weighted fil-

ter which is commonly used in the measurement of noise.

The shape of all weighting filters is similar, with the peak of

the curve usually occurring in the 3 kHz–7 kHz region as

shown below.

SUPPLY BYPASSING

The LM3876 has excellent power supply rejection and does

not require a regulated supply. However, to eliminate pos-

sible oscillations all op amps and power op amps should

have their supply leads bypassed with low-inductance ca-

pacitors having short leads and located close to the package

terminals. Inadequate power supply bypassing will manifest

itself by a low frequency oscillation known as “motorboating”

or by high frequency instabilities. These instabilities can be

eliminated through multiple bypassing utilizing a large tanta-

lum or electrolytic capacitor (10 µF or larger) which is used to

absorb low frequency variations and a small ceramic capaci-

tor (0.1 µF) to prevent any high frequency feedback through

the power supply lines.

If adequate bypassing is not provided the current in the sup-

ply leads which is a rectified component of the load current

may be fed back into internal circuitry. This signal causes low

distortion at high frequencies requiring that the supplies be

bypassed at the package terminals with an electrolytic ca-

pacitor of 470 µF or more.

LEAD INDUCTANCE

Power op amps are sensitive to inductance in the output

lead, particularly with heavy capacitive loading. Feedback to

the input should be taken directly from the output terminal,

minimizing common inductance with the load.

Lead inductance can also cause voltage surges on the sup-

plies. With long leads to the power supply, energy is stored in

the lead inductance when the output is shorted. This energy

can be dumped back into the supply bypass capacitors when

the short is removed. The magnitude of this transient is re-

duced by increasing the size of the bypass capacitor near

the IC. With at least a 20 µF local bypass, these voltage

surges are important only if the lead length exceeds a couple

feet (>1 µH lead inductance). Twisting together the supply

and ground leads minimizes the effect.

DS011832-13

DS011832-14

www.national.com13

Application Information (Continued)

LAYOUT, GROUND LOOPS AND STABILITY

The LM3876 is designed to be stable when operated at a

closed-loop gain of 10 or greater, but as with any other

high-current amplifier, the LM3876 can be made to oscillate

under certain conditions. These usually involve printed cir-

cuit board layout or output/input coupling.

When designing a layout, it is important to return the load

ground, the output compensation ground, and the low level

(feedback and input) grounds to the circuit board common

ground point through separate paths. Otherwise, large cur-

rents flowing along a ground conductor will generate volt-

ages on the conductor which can effectively act as signals at

the input, resulting in high frequency oscillation or excessive

distortion. It is advisable to keep the output compensation

components and the 0.1 µF supply decoupling capacitors as

close as possible to the LM3876 to reduce the effects of PCB

trace resistance and inductance. For the same reason, the

ground return paths should be as short as possible.

In general, with fast, high-current circuitry, all sorts of prob-

lems can arise from improper grounding which again can be

avoided by returning all grounds separately to a common

point. Without isolating the ground signals and returning the

grounds to a common point, ground loops may occur.

“Ground Loop” is the term used to describe situations occur-

ring in ground systems where a difference in potential exists

between two ground points. Ideally a ground is a ground, but

unfortunately, in order for this to be true, ground conductors

with zero resistance are necessary. Since real world ground

leads possess finite resistance, currents running through

them will cause finite voltage drops to exist. If two ground re-

turn lines tie into the same path at different points there will

be a voltage drop between them. The first figure below

shows a common ground example where the positive input

ground and the load ground are returned to the supply

ground point via the same wire. The addition of the finite wire

resistance, R

, results in a voltage difference between the

two points as shown below.

The load current I

will be much larger than input bias current

, thus V

will follow the output voltage directly, i.e. in phase.

Therefore the voltage appearing at the non-inverting input is

effectively positive feedback and the circuit may oscillate. If

there were only one device to worry about then the values of

and R

would probably be small enough to be ignored;

however, several devices normally comprise a total system.

Any ground return of a separate device, whose output is in

phase, can feedback in a similar manner and cause instabili-

ties. Out of phase ground loops also are troublesome, caus-

ing unexpected gain and phase errors.

The solution to most ground loop problems is to always use

a single-point ground system, although this is sometimes im-

practical. The third figure below is an example of a

single-point ground system.

The single-point ground concept should be applied rigor-

ously to all components and all circuits when possible. Viola-

tions of single-point grounding are most common among

printed circuit board designs, since the circuit is surrounded

by large ground areas which invite the temptation to run a

device to the closest ground spot. As a final rule, make all

ground returns low resistance and low inductance by using

large wire and wide traces.

Occasionally, current in the output leads (which function as

antennas) can be coupled through the air to the amplifier in-

put, resulting in high-frequency oscillation. This normally

happens when the source impedance is high or the input

leads are long. The problem can be eliminated by placing a

small capacitor, C

, (on the order of 50 pF to 500 pF) across

the LM3876 input terminals. Refer to the External Compo-

nents Description section relating to component interaction

with C

REACTIVE LOADING

It is hard for most power amplifiers to drive highly capacitive

loads very effectively and normally results in oscillations or

ringing on the square wave response. If the output of the

LM3876 is connected directly to a capacitor with no series

resistance, the square wave response will exhibit ringing if

the capacitance is greater than about 0.2 µF. If highly capaci-

tive loads are expected due to long speaker cables, a

method commonly employed to protect amplifiers from low

impedances at high frequencies is to couple to the load

through a 10Ωresistor in parallel with a 0.7 µH inductor. The

inductor-resistor combination as shown in the Typical Appli-

cation Circuit isolates the feedback amplifier from the load

by providing high output impedance at high frequencies thus

allowing the 10Ωresistor to decouple the capacitive load and

reduce the Q of the series resonant circuit. The LR combina-

tion also provides low output impedance at low frequencies

thus shorting out the 10Ωresistor and allowing the amplifier

to drive the series RC load (large capacitive load due to long

speaker cables) directly.

GENERALIZED AUDIO POWER AMPLIFIER DESIGN

The system designer usually knows some of the following

parameters when starting an audio amplifier design:

Desired Power Output Input Level

Input Impedance Load Impedance

Maximum Supply Voltage Bandwidth

The power output and load impedance determine the power

supply requirements, however, depending upon the applica-

tion some system designers may be limited to certain maxi-

mum supply voltages. If the designer does have a power

supply limitation, he should choose a practical load imped-

DS011832-15

www.national.com 14

Application Information (Continued)

ance which would allow the amplifier to provide the desired

output power, keeping in mind the current limiting capabili-

ties of the device. In any case, the output signal swing and

current are found from (where P

is the average output

power):

(5)

(6)

To determine the maximum supply voltage the following pa-

rameters must be considered. Add the dropout voltage (5V

for LM3876) to the peak output swing, V

opeak

, to get the sup-

ply rail value (i.e. ±(V

opeak

+ Vod) at a current of I

opeak

). The

regulation of the supply determines the unloaded voltage,

usually about 15%higher. Supply voltage will also rise 10%

during high line conditions. Therefore, the maximum supply

voltage is obtained from the following equation:

Max. supplies ≈±(V

opeak

+ Vod)(1 + regulation)(1.1)(7)

The input sensitivity and the output power specs determine

the minimum required gain as depicted below:

(8)

Normally the gain is set between 20 and 200; for a 40W, 8Ω

audio amplifier this results in a sensitivity of 894 mV and

89 mV, respectively. Although higher gain amplifiers provide

greater output power and dynamic headroom capabilities,

there are certain shortcomings that go along with the so

called “gain.” The input referred noise floor is increased and

hence the SNR is worse. With the increase in gain, there is

also a reduction of the power bandwidth which results in a

decrease in feedback thus not allowing the amplifier to re-

spond quickly enough to nonlinearities. This decreased abil-

ity to respond to nonlinearities increases the THD + N speci-

fication.

The desired input impedance is set by R

. Very high values

can cause board layout problems and DC offsets at the out-

put. The value for the feedback resistance, R

, should be

chosen to be a relatively large value (10 kΩ–100 kΩ), and

the other feedback resistance, Ri, is calculated using stan-

dard op amp configuration gain equations. Most audio ampli-

fiers are designed from the non-inverting amplifier configura-

tion.

DESIGN A 40W/8ΩAUDIO AMPLIFIER

Given:

Power Output 40W

Load Impedance 8Ω

Input Level 1V(max)

Input Impedance 100 kΩ

Bandwidth 20 Hz–20 kHz ±0.25 dB

Equations (5), (6)

give:

40W/8ΩV

opeak

=25.3V I

opeak

=3.16A

Therefore the supply required is: ±30.3V @3.16A

With 15%regulation and high line the final supply voltage is

±38.3V using

Equation (7)

. At this point it is a good idea to

check the Power Output vs Supply Voltage to ensure that the

required output power is obtainable from the device while

maintaining low THD + N. It is also good to check the Power

Dissipation vs Supply Voltage to ensure that the device can

handle the internal power dissipation. At the same time de-

signing in a relatively practical sized heat sink with a low

thermal resistance is also important. Refer to Typical Per-

formance Characteristics graphs and the Thermal Con-

siderations section for more information.

The minimum gain from

Equation (8)

is: A

≥18

We select a gain of 21 (Non-Inverting Amplifier); resulting in

a sensitivity of 894 mV.

Letting R

equal 100 kΩgives the required input imped-

ance, however, this would eliminate the “volume control” un-

less an additional input impedance was placed in series with

the 10 kΩpotentiometer that is depicted in

Figure 1

. Adding

the additional 100 kΩresistor would ensure the minumum

required input impedance.

For low DC offsets at the output we let R

=100 kΩ. Solving

for Ri (Non-Inverting Amplifier) gives the following:

Ri =R

/(A

−1)=100k/(21 − 1) =5kΩ; use 5.1 kΩ

The bandwidth requirement must be stated as a pole, i.e.,

the 3 dB frequency. Five times away from a pole gives

0.17 dB down, which is better than the required 0.25 dB.

Therefore: f

=20 Hz/5 =4Hz

=20 kHz x 5 =100 kHz

At this point, it is a good idea to ensure that the

Gain-Bandwidth Product for the part will provide the de-

signed gain out to the upper 3 dB point of 100 kHz. This is

why the minimum GBWP of the LM3876 is important.

GBWP ≥A

xf3dB=21 x 100 kHz =2.1 MHz

GBWP =2.0 MHz (min) for the LM3876

Solving for the low frequency roll-off capacitor, Ci, we have:

Ci ≥1/(2πRi f

)=7.8 µF; use 10 µF.

Definition of Terms

Input Offset Voltage: The absolute value of the voltage

which must be applied between the input terminals through

two equal resistances to obtain zero output voltage and cur-

rent.

Input Bias Current: The absolute value of the average of

the two input currents with the output voltage and current at

zero.

Input Offset Current: The absolute value of the difference

in the two input currents with the output voltage and current

at zero.

Input Common-Mode Voltage Range (or Input Voltage

Range): The range of voltages on the input terminals for

which the amplifier is operational. Note that the specifica-

tions are not guaranteed over the full common-mode voltage

range unless specifically stated.

Common-Mode Rejection: The ratio of the input

common-mode voltage range to the peak-to-peak change in

input offset voltage over this range.

Power Supply Rejection: The ratio of the change in input

offset voltage to the change in power supply voltages pro-

ducing it.

Quiescent Supply Current: The current required from the

power supply to operate the amplifier with no load and the

output voltage and current at zero.

Slew Rate: The internally limited rate of change in output

voltage with a large amplitude step function applied to the in-

put.

www.national.com15

Definition of Terms (Continued)

Class B Amplifier: The most common type of audio power

amplifier that consists of two output devices each of which

conducts for 180˚ of the input cycle. The LM3876 is a

Quasi-AB type amplifier.

Crossover Distortion: Distortion caused in the output stage

of a class B amplifier. It can result from inadequate bias cur-

rent providing a dead zone where the output does not re-

spond to the input as the input cycle goes through its zero

crossing point. Also for ICs an inadequate frequency re-

sponse of the output PNP device can cause a turn-on delay

giving crossover distortion on the negative going transition

through zero crossing at the higher audio frequencies.

THD+N:Total Harmonic Distortion plus Noise refers to the

measurement technique in which the fundamental compo-

nent is removed by a bandreject (notch) filter and all remain-

ing energy is measured including harmonics and noise.

Signal-to-Noise Ratio: The ratio of a system’s output signal

level to the system’s output noise level obtained in the ab-

sence of a signal. The output reference signal is either speci-

fied or measured at a specified distortion level.

Continuous Average Output Power: The minimum sine

wave continuous average power output in watts (or dBW)

that can be delivered into the rated load, over the rated

bandwidth, at the rated maximum total harmonic distortion.

Music Power: A measurement of the peak output power ca-

pability of an amplifier with either a signal duration suffi-

ciently short that the amplifier power supply does not sag

during the measurement, or when high quality external

power supplies are used. This measurement (an IHF stan-

dard) assumes that with normal music program material the

amplifier power supplies will sag insignificantly.

Peak Power: Most commonly referred to as the power out-

put capability of an amplifier that can be delivered to the

load; specified by the part’s maximum voltage swing.

Headroom: The margin between an actual signal operating

level (usually the power rating of the amplifier with particular

supply voltages, a rated load value, and a rated THD + N fig-

ure) and the level just before clipping distortion occurs, ex-

pressed in decibels.

Large Signal Voltage Gain: The ratio of the output voltage

swing to the differential input voltage required to drive the

output from zero to either swing limit. The output swing limit

is the supply voltage less a specified quasi-saturation volt-

age.A pulse of short enough duration to minimize thermal ef-

fects is used as a measurement signal.

Output-Current Limit: The output current with a fixed out-

put voltage and a large input overdrive. The limiting current

drops with time once SPiKe protection circuitry is activated.

Output Saturation Threshold (Clipping Point): The output

swing limit for a specified input drive beyond that required for

zero output. It is measured with respect to the supply to

which the output is swinging.

Output Resistance: The ratio of the change in output volt-

age to the change in output current with the output around

zero.

Power Dissipation Rating: The power that can be dissi-

pated for a specified time interval without activating the pro-

tection circuitry. For time intervals in excess of 100 ms, dis-

sipation capability is determined by heat sinking of the IC

package rather than by the IC itself.

Thermal Resistance: The peak, junction-temperature rise,

per unit of internal power dissipation (units in ˚C/W), above

the case temperature as measured at the center of the pack-

age bottom.

The DC thermal resistance applies when one output transis-

tor is operating continuously. The AC thermal resistance ap-

plies with the output transistors conducting alternately at a

high enough frequency that the peak capability of neither

transistor is exceeded.

Power Bandwidth: The power bandwidth of an audio ampli-

fier is the frequency range over which the amplifier voltage

gain does not fall below 0.707 of the flat band voltage gain

specified for a given load and output power.

Power bandwidth also can be measured by the frequencies

at which a specified level of distortion is obtained while the

amplifier delivers a power output 3 dB below the rated out-

put. For example, an amplifier rated at 60W with ≤0.25%

THD + N, would make its power bandwidth measured as the

difference between the upper and lower frequencies at which

0.25%distortion was obtained while the amplifier was deliv-

ering 30W.

Gain-Bandwidth Product: The Gain-Bandwidth Product is

a way of predicting the high-frequency usefulness of an op

amp. The Gain-Bandwidth Product is sometimes called the

unity-gain frequency or unity-gain cross frequency because

the open-loop gain characteristic passes through or crosses

unity gain at this frequency. Simply, we have the following re-

lationship: A

CL1

CL2

Assuming that at unity-gain (A

CL1

=1 or (0 dB)) fu =fi =

GBWP, then we have the following: GBWP =A

CL2

xf2

This says that once fu (GBWP) is known for an amplifier,

then the open-loop gain can be found at any frequency. This

is also an excellent equation to determine the 3 dB point of a

closed-loop gain, assuming that you know the GBWP of the

device. Refer to the diagram on the following page.

Biamplification: The technique of splitting the audio fre-

quency spectrum into two sections and using individual

power amplifiers to drive a separate woofer and tweeter.

Crossover frequencies for the amplifiers usually vary be-

tween 500 Hz and 1600 Hz. “Biamping” has the advantages

of allowing smaller power amps to produce a given sound

pressure level and reducing distortion effects prodused by

overdrive in one part of the frequency spectrum affecting the

other part.

C.C.I.R./A.R.M.:

Literally: International Radio Consultative Committee

Average Responding Meter

This refers to a weighted noise measurement for a Dolby B

type noise reduction system. A filter characteristic is used

that gives a closer correlation of the measurement with the

subjective annoyance of noise to the ear. Measurements

made with this filter cannot necessarily be related to un-

weighted noise measurements by some fixed conversion

factor since the answers obtained will depend on the spec-

trum of the noise source.

S.P.L.: Sound Pressure Level—usually measured with a

microphone/meter combination calibrated to a pressure level

of 0.0002 µBars (approximately the threshold hearing level).

S.P.L. =20 Log 10P/0.0002 dB

where P is the R.M.S. sound pressure in microbars.

(1 Bar =1 atmosphere =14.5 lb/in

=194 dB S.P.L.).

www.national.com 16

Definition of Terms (Continued)

DS011832-16

www.national.com17

Physical Dimensions inches (millimeters) unless otherwise noted

Order Number LM3876T

NS Package Number TA11B

Order Number LM3876TF

NS Package Number TF11B

www.national.com 18

Notes

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COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein:

1. Life support devices or systems are devices or

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into the body, or (b) support or sustain life, and

whose failure to perform when properly used in

accordance with instructions for use provided in the

labeling, can be reasonably expected to result in a

significant injury to the user.

2. A critical component is any component of a life

support device or system whose failure to perform

can be reasonably expected to cause the failure of

the life support device or system, or to affect its

safety or effectiveness.

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www.national.com

LM3876 Overture Audio Power Amplifier Series

High-Performance 56W Audio Power Amplifier w/Mute

National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications.