AD7714ARZ-5REEL - ANALOG DEVICES

AD7714

FUNCTIONAL BLOCK DIAGRAM

FEATURES

Charge Balancing ADC

24 Bits No Missing Codes

0.0015% Nonlinearity

Five-Channel Programmable Gain Front End

Gains from 1 to 128

Can Be Configured as Three Fully Differential

Inputs or Five Pseudo-Differential Inputs

Three-Wire Serial Interface

SPI™, QSPI™, MICROWIRE™ and DSP Compatible

3 V (AD7714-3) or 5 V (AD7714-5) Operation

Low Noise (<150 nV rms)

Low Current (350␣ ␮A typ) with Power-Down (5 ␮A typ)

AD7714Y Grade:

+2.7 V to 3.3 V or +4.75 V to +5.25 V Operation

0.0010% Linearity Error

–40ⴗC to +105ⴗC Temperature Range

Schmitt Trigger on SCLK and DIN

Low Current (226␣ ␮A typ) with Power-Down (4 ␮A typ)

Lower Power Dissipation than Standard AD7714

Available in 24-Lead TSSOP Package

Low-Pass Filter with Programmable Filter Cutoffs

Ability to Read/Write Calibration Coefficients

APPLICATIONS

Portable Industrial Instruments

Portable Weigh Scales

Loop-Powered Systems

Pressure Transducers

3 V/5 V, CMOS, 500 ␮A

Signal Conditioning ADC

GENERAL DESCRIPTION†

The AD7714 is a complete analog front end for low-frequency

measurement applications. The device accepts low level signals

directly from a transducer and outputs a serial digital word. It

employs a sigma-delta conversion technique to realize up to 24

bits of no missing codes performance. The input signal is applied

to a proprietary programmable gain front end based around an

analog modulator. The modulator output is processed by an on-

chip digital filter. The first notch of this digital filter can be

programmed via the on-chip control register allowing adjust-

ment of the filter cutoff and settling time.

The part features three differential analog inputs (which can also

be configured as five pseudo-differential analog inputs) as well as a

differential reference input. It operates from a single supply (+3␣ V

or +5␣ V). The AD7714 thus performs all signal conditioning and

conversion for a system consisting of up to five channels.

The AD7714 is ideal for use in smart, microcontroller- or DSP-

based systems. It features a serial interface that can be configured

REV. C

Information furnished by Analog Devices is believed to be accurate and

reliable. However, no responsibility is assumed by Analog Devices for its

use, nor for any infringements of patents or other rights of third parties

which may result from its use. No license is granted by implication or

otherwise under any patent or patent rights of Analog Devices.

for three-wire operation. Gain settings, signal polarity and channel

selection can be configured in software using the serial port. The

AD7714 provides self-calibration, system calibration and back-

ground calibration options and also allows the user to read and

write the on-chip calibration registers.

CMOS construction ensures very low power dissipation, and the

power-down mode reduces the standby power consumption to

15␣ µW typ. The part is available in a 24-pin, 0.3 inch-wide, plastic

dual-in-line package (DIP); a 24-lead small outline (SOIC)

package, a 28-lead shrink small outline package (SSOP) and a

24-lead thin shrink small outline package (TSSOP).

PRODUCT HIGHLIGHTS

1. The AD7714Y offers the following features in addition to the

standard AD7714: wider temperature range, Schmitt trigger

on SCLK and DIN, operation down to 2.7 V, lower power

consumption, better linearity, and availability in 24-lead

TSSOP package.

2. The AD7714 consumes less than 500 µA (f

CLK IN

= 1␣ MHz)

or 1 mA (f

CLK IN

= 2.5␣ MHz) in total supply current, making

it ideal for use in loop-powered systems.

3. The programmable gain channels allow the AD7714 to ac-

cept input signals directly from a strain gage or transducer

removing a considerable amount of signal conditioning.

4. The AD7714 is ideal for microcontroller or DSP processor

applications with a three-wire serial interface reducing the num-

ber of interconnect lines and reducing the number of opto-

couplers required in isolated systems. The part contains

on-chip registers that allow control over filter cutoff, input gain,

channel selection, signal polarity and calibration modes.

5. The part features excellent static performance specifications

with 24-bit no missing codes, ±0.0015% accuracy and low

rms noise (140 nV). Endpoint errors and the effects of tem-

perature drift are eliminated by on-chip self-calibration,

which removes zero-scale and full-scale errors.

†See page 39 for data sheet index.

SPI and QSPI are trademarks of Motorola, Inc.

MICROWIRE is a trademark of National Semiconductor Corporation.

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.

Tel: 781/329-4700 World Wide Web Site: http://www.analog.com

Fax: 781/326-8703 © Analog Devices, Inc., 1998

REF IN(+)

MCLK IN

MCLK OUT

A = 1–128

CHARGE

BALANCING

A/D CONVERTER

REF IN(–)

AD7714

BUFFER

AGND

BUFFER

SERIAL INTERFACE

AGND DGND

Σ-∆

MODULATOR SYNC

STANDBY

DIGITAL FILTER

DVDD

AVDD

PGA

1mA

AIN1

AIN2

AIN3

AIN4

AIN5

AIN6

SWITCHING

MATRIX

CLOCK

GENERATION

SCLK

DIN

DOUT

POL DRDY RESET

Parameter A Versions

Units Conditions/Comments

STATIC PERFORMANCE

No Missing Codes 24 Bits min Guaranteed by Design. Bipolar Mode. For Filter Notches ≤ 60 Hz

22 Bits min For Filter Notch = 100 Hz

18 Bits min For Filter Notch = 250 Hz

15 Bits min For Filter Notch = 500 Hz

12 Bits min For Filter Notch = 1 kHz

Output Noise See Tables I to IV Depends on Filter Cutoffs and Selected Gain

Integral Nonlinearity ±0.0015 % of FSR max Filter Notches ≤ 60 Hz

Unipolar Offset Error See Note 2

Unipolar Offset Drift

0.5 µV/°C typ For Gains of 1, 2, 4

0.3 µV/°C typ For Gains of 8, 16, 32, 64, 128

Bipolar Zero Error See Note 2

Bipolar Zero Drift

0.5 µV/°C typ For Gains of 1, 2, 4

0.3 µV/°C typ For Gains of 8, 16, 32, 64, 128

Positive Full-Scale Error

See Note 2

Full-Scale Drift

3, 5

0.5 µV/°C typ For Gains of 1, 2, 4

0.3 µV/°C typ For Gains of 8, 16, 32, 64, 128

Gain Error

See Note 2

Gain Drift

3, 7

0.5 ppm of FSR/°C typ

Bipolar Negative Full-Scale Error ±0.0015 % of FSR max Typically ±0.0004%

Bipolar Negative Full-Scale Drift

1µV/°C typ For Gains of 1, 2, 4

0.6 µV/°C typ For Gains of 8, 16, 32, 64, 128

ANALOG INPUTS/REFERENCE INPUTS Specifications for AIN and REF IN Unless Noted

Input Common-Mode Rejection (CMR) 90 dB min At DC. Typically 102 dB

Normal-Mode 50 Hz Rejection

100 dB min For Filter Notches of 10

, 25

, 50 Hz, ±0.02 × f

NOTCH

Normal-Mode 60 Hz Rejection

100 dB min For Filter Notches of 10

, 30

, 60 Hz, ±0.02 × f

NOTCH

Common-Mode 50 Hz Rejection

150 dB min For Filter Notches of 10

, 25

, 50 Hz, ±0.02 × f

NOTCH

Common-Mode 60 Hz Rejection

150 dB min For Filter Notches of 10

, 30

, 60 Hz, ±0.02 × f

NOTCH

Common-Mode Voltage Range

AGND to AV

V min to V max AIN for BUFFER = 0 and REF IN

Absolute AIN/REF IN Voltage

AGND – 30 mV V min AIN for BUFFER = 0 and REF IN

+ 30 mV V max

Absolute/Common-Mode AIN Voltage

AGND + 50 mV V min BUFFER = 1. A Version

– 1.5 V V max

AIN Input Current

1 nA max A Version

AIN Sampling Capacitance

7 pF max

AIN Differential Voltage Range

0 to +V

REF

/GAIN

nom Unipolar Input Range (B/U Bit of Filter High Register = 1)

±V

REF

/GAIN nom Bipolar Input Range (B/U Bit of Filter High Register = 0)

AIN Input Sampling Rate, f

GAIN × f

CLK␣ IN

/64 For Gains of 1, 2, 4

CLK␣ IN

/8 For Gains of 8, 16, 32, 64, 128

REF IN(+) – REF IN(–) Voltage +2.5 V nom ±1% for Specified Performance. Functional with Lower V

REF

REF IN Input Sampling Rate, f

CLK IN

/64

LOGIC INPUTS

Input Current ±10 µA max

All Inputs Except MCLK IN

INL

, Input Low Voltage 0.8 V max DV

= +5 V

INL

, Input Low Voltage 0.4 V max DV

= +3.3␣ V

INH

, Input High Voltage 2.4 V min DV

= +5 V

INH

, Input High Voltage 2.0 V min DV

= +3.3 V

MCLK IN Only

INL

, Input Low Voltage 0.8 V max DV

= +5␣ V

INL

, Input Low Voltage 0.4 V max DV

= +3.3␣ V

INH

, Input High Voltage 3.5 V min DV

= +5␣ V

INH

, Input High Voltage 2.5 V min DV

= +3.3␣ V

LOGIC OUTPUTS (Including MCLK OUT)

, Output Low Voltage 0.4 V max I

SINK

= 800␣ µA Except for MCLK OUT.

= +5 V

, Output Low Voltage 0.4 V max I

SINK

= 100␣ µA Except for MCLK OUT.

= +3.3 V

, Output High Voltage 4.0 V min I

SOURCE

= 200 µA Except for MCLK OUT.

= +5␣ V

, Output High Voltage DV

– 0.6 V V min I

SOURCE

= 100 µA Except for MCLK OUT.

= +3.3␣ V

Floating State Leakage Current ±10 µA max

Floating State Output Capacitance

9 pF typ

Data Output Coding Binary Unipolar Mode

Offset Binary Bipolar Mode

NOTES

Temperature range is as follows: A Versions: –40°C to +85°C.

A calibration is effectively a conversion so these errors will be of the order of the conversion noise shown in Tables I to IV. This applies after calibration at the temperature of interest.

Recalibration at any temperature will remove these drift errors.

Positive Full-Scale Error includes Zero-Scale Errors (Unipolar Offset Error or Bipolar Zero Error) and applies to both unipolar and bipolar input ranges.

Full-Scale Drift includes Zero-Scale Drift (Unipolar Offset Drift or Bipolar Zero Drift) and applies to both unipolar and bipolar input ranges.

Gain Error does not include Zero-Scale Errors. It is calculated as Full-Scale Error—Unipolar Offset Error for unipolar ranges and Full-Scale Error—Bipolar Zero Error for

bipolar ranges.

AD7714-5–SPECIFICATIONS

(AVDD = +5␣ V, DVDD = +3.3␣ V or +5␣ V, REF IN(+) = +2.5␣ V; REF␣ IN(–) = AGND;

fCLK IN = 2.4576␣ MHz unless otherwise noted. All specifications TMIN to TMAX unless otherwise noted.)

REV. C

–2–

Parameter A Versions Units Conditions/Comments

STATIC PERFORMANCE

No Missing Codes 24 Bits min Guaranteed by Design. Bipolar Mode. For Filter Notches ≤ 60 Hz

22 Bits min For Filter Notch = 100 Hz

18 Bits min For Filter Notch = 250 Hz

15 Bits min For Filter Notch = 500 Hz

12 Bits min For Filter Notch = 1 kHz

Output Noise See Tables I to IV Depends on Filter Cutoffs and Selected Gain

Integral Nonlinearity ±0.0015 % of FSR max Filter Notches ≤ 60 Hz

Unipolar Offset Error See Note 2

Unipolar Offset Drift

0.4 µV/°C typ For Gains of 1, 2, 4

0.1 µV/°C typ For Gains of 8, 16, 32, 64, 128

Bipolar Zero Error See Note 2

Bipolar Zero Drift

0.4 µV/°C typ For Gains of 1, 2, 4

0.1 µV/°C typ For Gains of 8, 16, 32, 64, 128

Positive Full-Scale Error

See Note 2

Full-Scale Drift

3, 5

0.4 µV/°C typ For Gains of 1, 2, 4

0.1 µV/°C typ For Gains of 8, 16, 32, 64, 128

Gain Error

See Note 2

Gain Drift

3, 7

0.2 ppm of FSR/°C typ

Bipolar Negative Full-Scale Error ±0.003 % of FSR max Typically ±0.0004%

Bipolar Negative Full-Scale Drift

1µV/°C typ For Gains of 1, 2, 4

0.6 µV/°C typ For Gains of 8, 16, 32, 64, 128

ANALOG INPUTS/REFERENCE INPUTS Specifications for AIN and REF IN Unless Noted

Input Common-Mode Rejection (CMR) 90 dB min At DC. Typically 102 dB.

Normal-Mode 50 Hz Rejection

100 dB min For Filter Notches of 10 Hz, 25 Hz, 50 Hz, ±0.02 × f

NOTCH

Normal-Mode 60 Hz Rejection

100 dB min For Filter Notches of 10 Hz, 30 Hz, 60 Hz, ±0.02 × f

NOTCH

Common-Mode 50 Hz Rejection

150 dB min For Filter Notches of 10 Hz, 25 Hz, 50 Hz, ±0.02 × f

NOTCH

Common-Mode 60 Hz Rejection

150 dB min For Filter Notches of 10 Hz, 30 Hz, 60 Hz, ±0.02 × f

NOTCH

Common-Mode Voltage Range

AGND to AV

V min to V max AIN for BUFFER = 0 and REF IN

Absolute AIN/REF IN Voltage

AGND – 30 mV V min AIN for BUFFER = 0 and REF IN

+ 30 mV V max

Absolute/Common-Mode AIN Voltage

AGND + 50 mV V min BUFFER = 1

– 1.5 V V max

AIN Input Current

1 nA max

AIN Sampling Capacitance

7 pF max

AIN Differential Voltage Range

0 to +V

REF

/GAIN

nom Unipolar Input Range (B/U Bit of Filter High Register = 1)

±V

REF

/GAIN nom Bipolar Input Range (B/U Bit of Filter High Register = 0)

AIN Input Sampling Rate, f

GAIN × f

CLK␣ IN

/64 For Gains of 1, 2, 4

CLK␣ IN

/8 For Gains of 8, 16, 32, 64, 128

REF IN(+) – REF IN(–) Voltage +1.25 V nom ±1% for Specified Performance. Part Functions with

Lower V

REF

REF IN Input Sampling Rate, f

CLK IN

/64

LOGIC INPUTS

Input Current ±10 µA max

All Inputs Except MCLK IN

INL

, Input Low Voltage 0.4 V max

INH

, Input High Voltage 2.0 V min

MCLK IN Only

INL

, Input Low Voltage 0.4 V max

INH

, Input High Voltage 2.5 V min

LOGIC OUTPUTS (Including MCLK OUT)

, Output Low Voltage 0.4 V max I

SINK

= 100␣ µA Except for MCLK OUT

, Output High Voltage DV

– 0.6 V min I

SOURCE

= 100 µA Except for MCLK OUT

Floating State Leakage Current ±10 µA max

Floating State Output Capacitance

9 pF typ

Data Output Coding Binary Unipolar Mode

Offset Binary Bipolar Mode

NOTES

Gain Error Drift does not include Unipolar Offset Drift/Bipolar Zero Drift. It is effectively the drift of the part if zero-scale calibrations only were performed as is the case with

background calibration.

These numbers are guaranteed by design and/or characterization.

The common-mode voltage range on the input pairs applies provided the absolute input voltage specification is obeyed.

The input voltage range on the analog inputs is given here with respect to the voltage on the respective negative input of its differential or pseudo-differential pair. See Table VII

for which inputs form differential pairs.

REF

= REF IN(+) – REF IN(–).

These logic output levels apply to the MCLK OUT output only when it is loaded with a single CMOS load.

Sample tested at +25°C to ensure compliance.

See Burnout Current section.

AD7714-3–SPECIFICATIONS

(AVDD = +3.3␣ V, DVDD = +3.3␣ V, REF IN(+) = +1.25␣ V; REF␣ IN(–) = AGND;

fCLK IN = 2.4576␣ MHz unless otherwise noted. All specifications TMIN to TMAX unless otherwise noted.)

AD7714

REV. C –3–

AD7714–SPECIFICATIONS

Parameter A Versions Units Conditions/Comments

TRANSDUCER BURNOUT

Current 1 µA nom

Initial Tolerance ±10 % typ

Drift 0.1 %/°C typ

SYSTEM CALIBRATION

Positive Full-Scale Calibration Limit

(1.05 × V

REF

)/GAIN V max GAIN Is the Selected PGA Gain (Between 1 and 128)

Negative Full-Scale Calibration Limit

–(1.05 × V

REF

)/GAIN V max GAIN Is the Selected PGA Gain (Between 1 and 128)

Offset Calibration Limit

–(1.05 × V

REF

)/GAIN V max GAIN Is the Selected PGA Gain (Between 1 and 128)

Input Span

0.8 × V

REF

/GAIN V min GAIN Is the Selected PGA Gain (Between 1 and 128)

(2.1 × V

REF

)/GAIN V max GAIN Is the Selected PGA Gain (Between 1 and 128)

POWER REQUIREMENTS

Power Supply Voltages

Voltage (AD7714-3) +3 to +3.6 V For Specified Performance

Voltage (AD7714-5) +4.75 to +5.25 V For Specified Performance

Voltage +3 to +5.25 V For Specified Performance

Power Supply Currents

Current AV

= 3.3␣ V or 5␣ V. BST Bit of Filter High Register = 0

0.27 mA max Typically 0.2 mA. BUFFER = 0 V. f

CLK IN

= 1␣ MHz or 2.4576␣ MHz

0.6 mA max Typically 0.4 mA. BUFFER = DV

. f

CLK IN

= 1␣ MHz or 2.4576␣ MHz

= 3.3␣ V or 5␣ V. BST Bit of Filter High Register = 1

0.5 mA max Typically 0.3␣ mA. BUFFER = 0␣ V. f

CLK IN

= 2.4576␣ MHz

1.1 mA max Typically 0.8␣ mA. BUFFER = DV

. f

CLK IN

= 2.4576␣ MHz

Current

Digital I/Ps = 0␣ V or DV

DD.

External MCLK IN

0.23 mA max Typically 0.15␣ mA. DV

= 3.3␣ V. f

CLK IN

= 1␣ MHz

0.4 mA max Typically 0.3␣ mA. DV

= 5␣ V. f

CLK IN

= 1␣ MHz

0.5 mA max Typically 0.4␣ mA. DV

= 3.3␣ V. f

CLK IN

= 2.4576␣ MHz

0.8 mA max Typically 0.6␣ mA. DV

= 5␣ V. f

CLK IN

= 2.4576␣ MHz

Power Supply Rejection

See Note 20 dB typ

Normal-Mode Power Dissipation

= DV

= +3.3␣ V. Digital I/Ps = 0␣ V or DV

. External MCLK IN

1.65 mW max Typically 1.25␣ mW. BUFFER = 0␣ V. f

CLK IN

= 1␣ MHz. BST Bit = 0

2.75 mW max Typically 1.8␣ mW. BUFFER = +3.3␣ V. f

CLK IN

= 1␣ MHz. BST Bit = 0

2.55 mW max Typically 2␣ mW. BUFFER = 0␣ V. f

CLK IN

= 2.4576␣ MHz. BST Bit = 0

3.65 mW max Typically 2.6␣ mW. BUFFER = +3.3␣ V. f

CLK IN

= 2.4576␣ MHz. BST Bit = 0

Normal-Mode Power Dissipation AV

= DV

= +5␣ V. Digital I/Ps = 0␣ V or DV

. External MCLK IN

3.35 mW max Typically 2.5␣ mW. BUFFER = 0␣ V. f

CLK IN

= 1␣ MHz. BST Bit = 0

5 mW max Typically 3.5␣ mW. BUFFER = +5␣ V. f

CLK IN

= 1␣ MHz. BST Bit = 0

5.35 mW max Typically 4␣ mW. BUFFER = 0␣ V. f

CLK IN

= 2.4576␣ MHz. BST Bit = 0

7 mW max Typically 5␣ mW. BUFFER = +5␣ V. f

CLK IN

= 2.4576␣ MHz. BST Bit = 0

Standby (Power-Down) Current

40 µA max External MCLK IN = 0 V or DV

. Typically 20␣ µA. V

= +5 V

Standby (Power-Down) Current

10 µA max External MCLK IN = 0 V or DV

. Typically 5␣ µA. V

= +3.3 V

NOTES

After calibration, if the input voltage exceeds positive full scale, the converter will output all 1s. If the input is less than negative full scale, then the device outputs all 0s.

These calibration and span limits apply provided the absolute voltage on the analog inputs does not exceed AV

+ 30␣ mV or go more negative than AGND␣ –␣ 30␣ mV. The

offset calibration limit applies to both the unipolar zero point and the bipolar zero point.

For higher gains (≥8) at f

CLK␣ IN

= 2.4576␣ MHz, the BST bit of the Filter High Register must be set to 1. For other conditions, it can be set to 0.

When using a crystal or ceramic resonator across the MCLK pins as the clock source for the device, the DV

current and power dissipation will vary depending on the crystal

or resonator type (see Clocking and Oscillator Circuit section).

Measured at dc and applies in the selected passband. PSRR at 50 Hz will exceed 120 dB with filter notches of 5 Hz, 10 Hz, 25 Hz or 50 Hz. PSRR at 60 Hz will exceed 120 dB

with filter notches of 6 Hz, 10 Hz, 30 Hz or 60 Hz.

PSRR depends on gain. For Gain of 1 : 70 dB typ: For Gain of 2 : 75 dB typ; For Gain of 4 : 80 dB typ; For Gains of 8 to 128 : 85 dB typ.

If the external master clock continues to run in standby mode, the standby current increases to 150 µA typical with 5 V supplies and 75 µA typical with 3.3 V supplies. When

using a crystal or ceramic resonator across the MCLK pins as the clock source for the device, the internal oscillator continues to run in standby mode and the power dissipation

depends on the crystal or resonator type (see Standby Mode section).

Specifications subject to change without notice.

(AVDD = + 3.3␣ V to +5␣ V, DVDD = +3.3␣ V to +5␣ V, REF IN(+) = +1.25␣ V (AD7714-3) or +2.5␣ V

(AD7714-5); REF␣ IN(–) = AGND; MCLK␣ IN = 1␣ MHz to 2.4576␣ MHz unless otherwise noted. All specifications TMIN to TMAX unless otherwise noted.)

REV. C–4–

Parameter Y Versions

Units Conditions/Comments

STATIC PERFORMANCE

No Missing Codes 24 Bits min Guaranteed by Design. For Filter Notches ≤ 60 Hz

22 Bits min For Filter Notch = 100 Hz

18 Bits min For Filter Notch = 250 Hz

15 Bits min For Filter Notch = 500 Hz

12 Bits min For Filter Notch = 1 kHz

Output Noise See Tables I to IV Depends on Filter Cutoffs and Selected Gain

Integral Nonlinearity ±0.001 % of FSR max Filter Notches ≤ 60 Hz.

Unipolar Offset Error See Note 2

Unipolar Offset Drift

0.4 µV/°C typ For Gains of 1, 2, 4

0.1 µV/°C typ For Gains of 8, 16, 32, 64, 128

Bipolar Zero Error See Note 2

Bipolar Zero Drift

0.4 µV/°C typ For Gains of 1, 2, 4

0.1 µV/°C typ For Gains of 8, 16, 32, 64, 128

Positive Full-Scale Error

See Note 2

Full-Scale Drift

3, 5

0.4 µV/°C typ For Gains of 1, 2, 4

0.1 µV/°C typ For Gains of 8, 16, 32, 64, 128

Gain Error

See Note 2

Gain Drift

3, 7

0.2 ppm of FSR/

°C typ

Bipolar Negative Full-Scale Error

±0.0015 % of FSR max AV

= 5 V. Typically ±0.0004%

±0.003 % of FSR max AV

= 3 V. Typically ±0.0004%

Bipolar Negative Full-Scale Drift

1µV/°C typ For Gains of 1 to 4

0.6 µV/°C typ For Gains of 8 to 128

ANALOG INPUTS/REFERENCE INPUTS Specifications for AIN and REF IN Unless Noted

Input Common-Mode Rejection (CMR)

90 dB min At DC. Typically 102 dB.

Normal-Mode 50 Hz Rejection

100 dB min For Filter Notches of 10 Hz, 25 Hz, 50 Hz, ±0.02 × f

NOTCH

Normal-Mode 60 Hz Rejection

100 dB min For Filter Notches of 10 Hz, 30 Hz, 60 Hz, ±0.02 × f

NOTCH

Common-Mode 50 Hz Rejection

150 dB min For Filter Notches of 10 Hz, 25 Hz, 50 Hz, ±0.02 × f

NOTCH

Common-Mode 60 Hz Rejection

150 dB min For Filter Notches of 10 Hz, 30 Hz, 60 Hz, ±0.02 × f

NOTCH

Absolute/Common-Mode REF IN Voltage

AGND to AV

V min to V max

Absolute/Common-Mode AIN Voltage

AGND – 30 mV V min BUF Bit of Setup Register = 0

+ 30 mV V max

Absolute/Common-Mode AIN Voltage

8, 9

AGND + 50 mV V min BUF Bit of Setup Register = 1

– 1.5 V V max

AIN DC Input Current

1 nA max

AIN Sampling Capacitance

7 pF max

AIN Differential Voltage Range

0 to +V

REF

/GAIN

nom Unipolar Input Range (B/U Bit of Filter High Register = 1)

±V

REF

/GAIN nom Bipolar Input Range (B/U Bit of Filter High Register = 0)

AIN Input Sampling Rate, f

GAIN × f

CLK␣ IN

/64 For Gains of 1 to 4

CLK␣ IN

/8 For Gains of 8 to 128

Reference Input Range

REF IN(+) – REF IN(–) Voltage 1/1.75 V min/max AV

= 2.7 V to 3.3 V. V

REF

= 1.25 ±1% for Specified Performance

REF IN(+) – REF IN(–) Voltage 1/3.5 V min/max AV

= 4.75 V to 5.25 V. V

REF

= 2.5 ±1% for Specified Performance

REF IN Input Sampling Rate, f

CLK IN

/64

LOGIC INPUTS

Input Current ±10 µA max

All Inputs Except MCLK IN

INL

, Input Low Voltage 0.8 V max DV

= 5 V

0.4 V max DV

= 3 V

INH

, Input High Voltage 2.4 V min DV

= 5 V

2 V min DV

= 3 V

SCLK & DIN Only (Schmitt Triggered Input) DV

= 5 V NOMINAL

T+

1.4/3 V min/V max

T–

0.8/1.4 V min/V max

T+

– V

T–

0.4/0.8 V min/V max

SCLK & DIN Only (Schmitt Triggered Input) DV

= 3 V NOMINAL

T+

1/2.5 V min/V max

T–

0.4/1.1 V min/V max

T+

– V

T–

0.375/0.8 V min/V max

MCLK In Only DV

= 5 V NOMINAL

INL

, Input Low Voltage 0.8 V max

INH

, Input High Voltage 3.5 V min

MCLK In Only DV

= 3 V NOMINAL

INL

, Input Low Voltage 0.4 V max

INH

, Input High Voltage 2.5 V min

LOGIC OUTPUTS (Including MCLK OUT)

, Output Low Voltage 0.4 V max I

SINK

= 800␣ µA with DV

= 5 V. Except for MCLK OUT

, Output Low Voltage 0.4 V max I

SINK

= 100␣ µA with DV

= 3 V. Except for MCLK OUT

, Output High Voltage 4 V min I

SOURCE

= 200 µA with DV

= 5 V. Except for MCLK OUT

AD7714Y–SPECIFICATIONS

(AVDD = DVDD = +2.7␣ V to +3.3␣ V or 4.75 V to 5.25 V, REF IN(+) = +1.25␣ V; with AVDD = 3 V

and +2.5 V with AVDD = 5 V; REF␣ IN(–) = AGND; MCLK IN = 2.4576␣ MHz unless otherwise noted. All specifications TMIN to TMAX unless otherwise noted.)

AD7714

REV. C –5–

Parameter Y Versions Units Conditions/Comments

LOGIC OUTPUTS (Continued))

, Output High Voltage DV

– 0.6 V min I

SOURCE

= 100 µA with DV

= 3 V. Except for MCLK OUT

Floating State Leakage Current ±10 µA max

Floating State Output Capacitance

9 pF typ

ata Output Coding Binary Unipolar Mode

Offset Binary Bipolar Mode

TRANSDUCER BURNOUT

Current 1 µA nom

Initial Tolerance ±10 % typ

Drift 0.1 %/°C typ

SYSTEM CALIBRATION

Positive Full-Scale Calibration Limit

(1.05 × V

REF

)/GAIN V max GAIN Is the Selected PGA Gain (Between 1 and 128)

Negative Full-Scale Calibration Limit

–(1.05 × V

REF

)/GAIN V max GAIN Is the Selected PGA Gain (Between 1 and 128)

Offset Calibration Limit

–(1.05 × V

REF

)/GAIN V max GAIN Is the Selected PGA Gain (Between 1 and 128)

Input Span

0.8 × V

REF

/GAIN V min GAIN Is the Selected PGA Gain (Between 1 and 128)

(2.1 × V

REF

)/GAIN V max GAIN Is the Selected PGA Gain (Between 1 and 128)

POWER REQUIREMENTS

Power Supply Voltages

Voltage +2.7 to +3.3 or V

+4.75 to +5.25 V For Specified Performance

Voltage +2.7 to +5.25 V For Specified Performance

Power Supply Currents

Current AV

= 3 V or 5␣ V. BST Bit of Filter High Register = 0

, CLKDIS = 1

0.28 mA max Typically 0.22 mA. BUFFER = 0 V. f

CLK IN

= 1␣ MHz or 2.4576␣ MHz

0.6 mA max Typically 0.45 mA. BUFFER = DV

. f

CLK IN

= 1␣ MHz or 2.4576␣ MHz

= 3 V or 5␣ V. BST Bit of Filter High Register = 1

0.5 mA max Typically 0.38␣ mA. BUFFER = 0␣ V. f

CLK IN

= 2.4576␣ MHz

1.1 mA max Typically 0.8␣ mA. BUFFER = DV

. f

CLK IN

= 2.4576␣ MHz

Current

Digital I/Ps = 0␣ V or DV

DD.

External MCLK IN, CLKDIS = 1

0.080 mA max Typically 0.06␣ mA. DV

= 3 V. f

CLK IN

= 1␣ MHz

0.16 mA max Typically 0.13␣ mA. DV

= 5␣ V. f

CLK IN

= 1␣ MHz

0.18 mA max Typically 0.15␣ mA. DV

= 3 V. f

CLK IN

= 2.4576␣ MHz

0.35 mA max Typically 0.3 mA. DV

= 5␣ V. f

CLK IN

= 2.4576␣ MHz

Power Supply Rejection

See Note 20 dB typ

Normal-Mode Power Dissipation

= DV

= +3 V. Digital I/Ps = 0␣ V or DV

. External MCLK IN

BST Bit of Filter High Register = 0

1.05 mW max Typically 0.84␣ mW. BUFFER = 0␣ V. f

CLK IN

= 1␣ MHz. BST Bit = 0

2.04 mW max Typically 1.53␣ mW. BUFFER = +3 V. f

CLK IN

= 1␣ MHz. BST Bit = 0

1.35 mW max Typically 1.11␣ mW. BUFFER = 0␣ V. f

CLK IN

= 2.4576␣ MHz. BST Bit = 0

2.34 mW max Typically 1.9␣ mW. BUFFER = +3 V. f

CLK IN

= 2.4576␣ MHz. BST Bit = 0

Normal-Mode Power Dissipation AV

= DV

= +5␣ V. Digital I/Ps = 0␣ V or DV

. External MCLK IN

2.1 mW max Typically 1.75 mW. BUFFER = 0␣ V. f

CLK IN

= 1␣ MHz. BST Bit = 0

3.75 mW max Typically 2.9 mW. BUFFER = +5␣ V. f

CLK IN

= 1␣ MHz. BST Bit = 0

3.1 mW max Typically 2.6␣ mW. BUFFER = 0␣ V. f

CLK IN

= 2.4576␣ MHz. BST Bit = 0

4.75 mW max Typically 3.75␣ mW. BUFFER = +5␣ V. f

CLK IN

= 2.4576␣ MHz. BST Bit = 0

Standby (Power-Down) Current

18 µA max External MCLK IN = 0 V or DV

. Typically 9␣ µA. V

= +5 V

Standby (Power-Down) Current

10 µA max External MCLK IN = 0 V or DV

. Typically 4␣ µA. V

= +3 V

NOTES

Temperature range is as follows: Y Version: –40°C to +105°C.

A calibration is effectively a conversion so these errors will be of the order of the conversion noise shown in Tables I to IV. This applies after calibration at the temperature of interest.

Recalibration at any temperature will remove these drift errors.

Positive Full-Scale Error includes Zero-Scale Errors (Unipolar Offset Error or Bipolar Zero Error) and applies to both unipolar and bipolar input ranges.

Full-Scale Drift includes Zero-Scale Drift (Unipolar Offset Drift or Bipolar Zero Drift) and applies to both unipolar and bipolar input ranges.

Gain Error does not include Zero-Scale Errors. It is calculated as Full-Scale Error—Unipolar Offset Error for unipolar ranges and Full-Scale Error—Bipolar Zero Error for

bipolar ranges.

Gain Error Drift does not include Unipolar Offset Drift/Bipolar Zero Drift. It is effectively the drift of the part if zero-scale calibrations only were performed as is the case with background calibration.

These numbers are guaranteed by design and/or characterization.

The common-mode voltage range on the input pairs applies provided the absolute input voltage specification is obeyed.

The input voltage range on the analog inputs is given here with respect to the voltage on the respective negative input of its differential or pseudo-differential pair. See Table VII for which

inputs form differential pairs.

REF

= REF IN(+) – REF IN(–).

These logic output levels apply to the MCLK OUT output only when it is loaded with a single CMOS load.

Sample tested at +25°C to ensure compliance.

See Burnout Current section.

After calibration, if the input voltage exceeds positive full scale, the converter will output all 1s. If the input is less than negative full scale, then the device outputs all 0s.

These calibration and span limits apply provided the absolute voltage on the analog inputs does not exceed AV

+ 30␣ mV or go more negative than AGND␣ –␣ 30␣ mV. The offset calibration

limit applies to both the unipolar zero point and the bipolar zero point.

For higher gains (≥8) at f

CLK␣ IN

= 2.4576␣ MHz, the BST bit of the Filter High Register must be set to 1. For other conditions, it can be set to 0.

When using a crystal or ceramic resonator across the MCLK pins as the clock source for the device, the DV

current and power dissipation will vary depending on the crystal or resonator

type (see Clocking and Oscillator Circuit section).

Measured at dc and applies in the selected passband. PSRR at 50 Hz will exceed 120 dB with filter notches of 5 Hz, 10 Hz, 25 Hz or 50 Hz. PSRR at 60 Hz will exceed 120 dB with filter

notches of 6 Hz, 10 Hz, 30 Hz or 60 Hz.

PSRR depends on gain.

Gain 1 2 4 8–128

= 3 V 86 dB 78 dB 85 dB 93 dB

= 5 V 90 dB 78 dB 84 dB 91 dB

If the external master clock continues to run in standby mode, the standby current increases to 150 µA typical with 5 V supplies and 75 µA typical with 3.3 V supplies. When using a crystal

or ceramic resonator across the MCLK pins as the clock source for the device, the internal oscillator continues to run in standby mode and the power dissipation depends on the crystal or

resonator type (see Standby Mode section).

Specifications subject to change without notice.

AD7714Y

REV. C

–6–

AD7714

REV. C –7–

ORDERING GUIDE

Temperature Package

Model Supply Range Option*

AD7714AN-5 5 V –40°C to +85°C N-24

AD7714AR-5 5 V –40°C to +85°C R-24

AD7714ARS-5 5 V –40°C to +85°C RS-28

AD7714AN-3 3 V –40°C to +85°C N-24

AD7714AR-3 3 V –40°C to +85°C R-24

AD7714ARS-3 3 V –40°C to +85°C RS-28

AD7714YN 3 V/5 V –40°C to +105°C N-24

AD7714YR 3 V/5 V –40°C to +105°C R-24

AD7714YRU 3 V/5 V –40°C to +105°C RU-24

AD7714AChips-5 5 V –40°C to +85°CDie

AD7714AChips-3 3 V –40°C to +85°CDie

EVAL-AD7714-5EB 5 V Evaluation Board

EVAL-AD7714-3EB 3 V Evaluation Board

*N = Plastic DIP; R = SOIC; RS = SSOP; RU = Thin Shrink Small Outline.

TIMING CHARACTERISTICS

1, 2

(AVDD = DVDD = +2.7 V to +5.25 V; AGND = DGND = 0 V; fCLKIN = 2.5␣ MHz; Input Logic 0 = 0 V,

Logic 1 = DVDD unless otherwise noted.)

Limit at T

MIN

, T

MAX

Parameter (A, Y Versions) Units Conditions/Comments

CLKIN3, 4

400 kHz min Master Clock Frequency: Crystal/Resonator or Externally

Supplied

2.5 MHz max For Specified Performance

CLK IN LO

0.4 × t

CLK IN

ns min Master Clock Input Low Time. t

CLK IN

= 1/f

CLK IN

CLK IN HI

0.4 × t

CLK IN

ns min Master Clock Input High Time

DRDY

500 × t

CLK IN

ns nom DRDY High Time

100 ns min SYNC Pulsewidth

100 ns min RESET Pulsewidth

Read Operation

0 ns min DRDY to CS Setup Time

0 ns min CS Falling Edge to SCLK Active Edge Setup Time

0 ns min SCLK Active Edge to Data Valid Delay

80 ns max DV

= +5␣ V

100 ns max DV

= +3␣ V

100 ns min SCLK High Pulsewidth

100 ns min SCLK Low Pulsewidth

0 ns min CS Rising Edge to SCLK Active Edge Hold Time

10 ns min Bus Relinquish Time after SCLK Active Edge

60 ns max DV

= +5␣ V

100 ns max DV

= +3␣ V

100 ns max SCLK Active Edge to DRDY High

5, 8

Write Operation

0 ns min CS Falling Edge to SCLK Active Edge Setup Time

30 ns min Data Valid to SCLK Edge Setup Time

20 ns min Data Valid to SCLK Edge Hold Time

100 ns min SCLK High Pulsewidth

100 ns min SCLK Low Pulsewidth

0 ns min CS Rising Edge to SCLK Edge Hold Time

NOTES

Sample tested at +25°C to ensure compliance. All input signals are specified with tr = tf = 5 ns (10% to 90% of DV

) and timed from a voltage level of 1.6 V.

See Figures 6 and 7. Timing applies for all grades.

CLKIN Duty Cycle range is 45% to 55%. CLKIN must be supplied whenever the AD7714 is not in standby mode. If no clock is present in this case, the device can

draw higher current than specified and possibly become uncalibrated.

The AD7714 is production tested with f

CLKIN

at 2.4576␣ MHz (1␣ MHz for some I

tests). It is guaranteed by characterization to operate at 400␣ kHz.

SCLK active edge is falling edge of SCLK with POL = 1; SCLK active edge is rising edge of SCLK with POL = 0.

These numbers are measured with the load circuit of Figure 1 and defined as the time required for the output to cross the V

or V

limits.

These numbers are derived from the measured time taken by the data output to change 0.5␣ V when loaded with the circuit of Figure 1. The measured number is then

extrapolated back to remove effects of charging or discharging the 100 pF capacitor. This means that the times quoted in the timing characteristics are the true bus

relinquish times of the part and as such are independent of external bus loading capacitances.

DRDY returns high after the first read from the device after an output update. The same data can be read again, if required, while DRDY is high although care

should be taken that subsequent reads do not occur close to the next output update.

Specifications subject to change without notice.

Figure 1. Load Circuit for Access Time and Bus

Relinquish Time

TO OUTPUT

50pF

ISINK (800mA AT DVDD = +5V

100mA AT DVDD = +3.3V)

+1.6V

ISOURCE (200mA AT DVDD = +5V

100mA AT DVDD = +3.3V)

AD7714

REV. C–8–

DIP and SOIC/TSSOP

ABSOLUTE MAXIMUM RATINGS*

= +25°C unless otherwise noted)

to AGND . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +7 V

to DGND . . . . . . . . . . . . . . . . . . . . . . . . –0.3␣ V to +7␣ V

to AGND . . . . . . . . . . . . . . . . . . . . . . . . –0.3␣ V to +7␣ V

to DGND . . . . . . . . . . . . . . . . . . . . . . . . –0.3␣ V to +7␣ V

Analog Input Voltage to AGND . . . . . –0.3 V to AV

+ 0.3␣ V

Reference Input Voltage to AGND . . . –0.3 V to AV

+ 0.3␣ V

Digital Input Voltage to DGND . . . . . –0.3 V to DV

+ 0.3 V

Digital Output Voltage to DGND . . . . –0.3 V to DV

+ 0.3 V

Operating Temperature Range

Commercial (A Version) . . . . . . . . . . . . . . . –40°C to +85°C

Extended (Y Version) . . . . . . . . . . . . . . . . . –40°C to +105°C

Storage Temperature Range . . . . . . . . . . . . . –65°C to +150°C

Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . .+150°C

Plastic DIP Package, Power Dissipation . . . . . . . . . . . 450 mW

Thermal Impedance . . . . . . . . . . . . . . . . . . . . . 105°C/W

Lead Temperature (Soldering, 10 sec) . . . . . . . . . . . .+260°C

SOIC Package, Power Dissipation . . . . . . . . . . . . . . . . 450 mW

Thermal Impedance . . . . . . . . . . . . . . . . . . . . . . 75°C/W

Lead Temperature, Soldering

Vapor Phase (60 sec) . . . . . . . . . . . . . . . . . . . . . . +215°C

Infrared (15 sec) . . . . . . . . . . . . . . . . . . . . . . . . . . +220°C

SSOP Package, Power Dissipation . . . . . . . . . . . . . . . . 450 mW

Thermal Impedance . . . . . . . . . . . . . . . . . . . . . 109°C/W

Lead Temperature, Soldering

Vapor Phase (60 sec) . . . . . . . . . . . . . . . . . . . . . . +215°C

Infrared (15 sec) . . . . . . . . . . . . . . . . . . . . . . . . . . +220°C

TSSOP Package, Power Dissipation . . . . . . . . . . . . . . 450 mW

Thermal Impedance . . . . . . . . . . . . . . . . . . . . . 128°C/W

Lead Temperature, Soldering

Vapor Phase (60 sec) . . . . . . . . . . . . . . . . . . . . . . +215°C

Infrared (15 sec) . . . . . . . . . . . . . . . . . . . . . . . . . . +220°C

*Stresses above those listed under Absolute Maximum Ratings may cause perma-

nent damage to the device. This is a stress rating only; functional operation of the

device at these or any other conditions above those listed in the operational

sections of this specification is not implied. Exposure to absolute maximum rating

conditions for extended periods may affect device reliability.

CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily

accumulate on the human body and test equipment and can discharge without detection.

Although these devices feature proprietary ESD protection circuitry, permanent damage may still

occur on these devices if they are subjected to high energy electrostatic discharges. Therefore,

proper ESD precautions are recommended to avoid performance degradation or loss of functionality.

PIN CONFIGURATIONS

SSOP

SCLK

MCLK IN

DGND

DVDD

SYNC

AIN1

DRDY

AGND

MCLK OUT

POL

DIN

DOUT

AIN2 AIN6

AIN3 AIN5

AIN4

STANDBY

AVDD

TOP VIEW

(Not to Scale)

AD7714

RESET

REF IN(+)

REF IN(–)

BUFFER

NC = NO CONNECT

SCLK

MCLK IN

DGND

DVDD

SYNC

RESET

DRDY

MCLK OUT

POL

DIN

DOUT

NC NC

AIN1 AGND

AIN2 AIN6

AIN3 AIN5

AIN4 REF IN(+)

STANDBY REF IN(–)

AVDD BUFFER

TOP VIEW

(Not to Scale)

AD7714

WARNING!

ESD SENSITIVE DEVICE

AD7714

REV. C –9–

PIN FUNCTION DESCRIPTION

DIP/SOIC PIN NUMBERS

No. Mnemonic Function

1 SCLK Serial Clock. Logic Input. An external serial clock is applied to this input to access serial data from the

AD7714. This serial clock can be a continuous clock with all data transmitted in a continuous train of pulses.

Alternatively, it can be a noncontinuous clock with the information being transmitted to the AD7714 in smaller

batches of data.

2 MCLK IN Master Clock signal for the device. This can be provided in the form of a crystal/resonator or external clock. A

crystal/resonator can be tied across the MCLK IN and MCLK OUT pins. Alternatively, the MCLK IN pin can

be driven with a CMOS-compatible clock and MCLK OUT left unconnected. The part is specified with clock

input frequencies of both 1 MHz and 2.4576 MHz.

3 MCLK OUT When the master clock for the device is a crystal/resonator, the crystal/resonator is connected between MCLK

IN and MCLK␣ OUT. If an external clock is applied to the MCLK IN, MCLK OUT provides an inverted clock

signal. This clock can be used to provide a clock source for external circuits.

4 POL Clock Polarity. Logic Input. With this input low, the first transition of the serial clock in a data transfer

operation is from a low to a high. In microcontroller applications, this means that the serial clock should idle

low between data transfers. With this input high, the first transition of the serial clock in a data transfer

operation is from a high to a low. In microcontroller applications, this means that the serial clock should idle

high between data transfers.

5SYNC Logic Input which allows for synchronization of the digital filters and analog modulators when using a number

of AD7714s. While SYNC is low, the nodes of the digital filter, the filter control logic and the calibration

control logic are reset and the analog modulator is also held in its reset state. SYNC does not affect the digital

interface and does not reset DRDY if it is low.

6RESET Logic Input. Active low input which resets the control logic, interface logic, digital filter and analog modulator

of the part to power-on status.

7 AIN1 Analog Input Channel 1. Programmable-gain analog input which can be used as a pseudo-differential input

when used with AIN6 or as the positive input of a differential analog input pair when used with AIN2 (see

Communications Register section).

8 AIN2 Analog Input Channel 2. Programmable-gain analog input which can be used as a pseudo-differential input

when used with AIN6 or as the negative input of a differential analog input pair when used with AIN1 (see

Communications Register section).

9 AIN3 Analog Input Channel 3. Programmable-gain analog input which can be used as a pseudo-differential input

when used with AIN6 or as the positive input of a differential analog input pair when used with AIN4 (see

Communications Register section).

10 AIN4 Analog Input Channel 4. Programmable-gain analog input which can be used as a pseudo-differential input

when used with AIN6 or as the negative input of a differential analog input pair when used with AIN3 (see

Communications Register section).

11 STANDBY Logic Input. Taking this pin low shuts down the analog and digital circuitry, reducing current consumption to

typically 5 µA.

12 AV

Analog Positive Supply Voltage, A Grade Versions: +3.3␣ V nominal (AD7714-3) or +5␣ V nominal (AD7714-5);

Y Grade Versions: 3 V or 5 V nominal.

13 BUFFER Buffer Option Select. Logic Input. With this input low, the on-chip buffer on the analog input (after the

multiplexer and before the analog modulator) is shorted out. With the buffer shorted out the current flowing in

the AV

line is reduced to 270 µA. With this input high, the on-chip buffer is in series with the analog input

allowing the inputs to handle higher source impedances.

14 REF IN(–) Reference Input. Negative input of the differential reference input to the AD7714. The REF IN(–) can lie

anywhere between AV

and AGND provided REF␣ IN(+) is greater than REF IN(–).

15 REF IN(+) Reference Input. Positive input of the differential reference input to the AD7714. The reference input is

differential with the provision that REF IN(+) must be greater than REF IN(–). REF IN(+) can lie anywhere

between AV

and AGND.

16 AIN5 Analog Input Channel 5. Programmable-gain analog input which is the positive input of a differential analog

input pair when used with AIN6 (see Communications Register section).

17 AIN6 Analog Input Channel 6. Reference point for AIN1 through AIN4 in pseudo-differential mode or as the

negative input of a differential input pair when used with AIN5 (see Communications Register section).

18 AGND Ground reference point for analog circuitry.

AD7714

REV. C–10–

PIN FUNCTION DESCRIPTION (Continued)

No. Mnemonic Function

19 CS Chip Select. Active low Logic Input used to select the AD7714. With this input hard-wired low, the AD7714

can operate in its three-wire interface mode with SCLK, DIN and DOUT used to interface to the device. CS

can be used to select the device in systems with more than one device on the serial bus or as a frame

synchronization signal in communicating with the AD7714.

20 DRDY Logic output. A logic low on this output indicates that a new output word is available from the AD7714 data

read has taken place, after an output update, the DRDY line will return high for 500 × t

CLK␣ IN

cycles prior to

the next output update. This gives an indication of when a read operation should not be attempted to avoid

reading from the data register as it is being updated. DRDY is also used to indicate when the AD7714 has

completed its on-chip calibration sequence.

21 DOUT Serial Data Output with serial data being read from the output shift register on the part. This output shift

selection registers or data register depending on the register selection bits of the Communications Register.

22 DIN Serial Data Input with serial data being written to the input shift register on the part. Data from this input shift

registers depending on the register selection bits of the Communications Register.

23 DV

Digital Supply Voltage, A Grade Versions: +3.3␣ V or +5 V nominal; Y Grade Versions: 3 V or 5 V nominal.

24 DGND Ground reference point for digital circuitry.

BIPOLAR NEGATIVE FULL-SCALE ERROR

This is the deviation of the first code transition from the ideal

AIN(+) voltage (AIN(–) – V

REF

/GAIN + 0.5␣ LSB) when operat-

ing in the bipolar mode.

POSITIVE FULL-SCALE OVERRANGE

Positive Full-Scale Overrange is the amount of overhead avail-

able to handle input voltages on AIN(+) input greater than

AIN(–) + V

REF

/GAIN (for example, noise peaks or excess volt-

ages due to system gain errors in system calibration routines)

without introducing errors due to overloading the analog modu-

lator or overflowing the digital filter.

NEGATIVE FULL-SCALE OVERRANGE

This is the amount of overhead available to handle voltages on

AIN(+) below AIN(–) – V

REF

/GAIN without overloading the

analog modulator or overflowing the digital filter. Note that the

analog input will accept negative voltage peaks even in the uni-

polar mode provided that AIN(+) is greater than AIN(–) and

greater than AGND – 30␣ mV.

OFFSET CALIBRATION RANGE

In the system calibration modes, the AD7714 calibrates its

offset with respect to the analog input. The Offset Calibration

Range specification defines the range of voltages that the

AD7714 can accept and still calibrate offset accurately.

FULL-SCALE CALIBRATION RANGE

This is the range of voltages that the AD7714 can accept in the

system calibration mode and still calibrate full scale correctly.

INPUT SPAN

In system calibration schemes, two voltages applied in sequence

to the AD7714’s analog input define the analog input range.

The input span specification defines the minimum and maxi-

mum input voltages from zero to full scale that the AD7714 can

accept and still calibrate gain accurately.

TERMINOLOGY*

INTEGRAL NONLINEARITY

This is the maximum deviation of any code from a straight line

passing through the endpoints of the transfer function. The end-

points of the transfer function are zero scale (not to be confused

with bipolar zero), a point 0.5 LSB below the first code transi-

tion (000 . . . 000 to 000 . . . 001) and full scale, a point

0.5 LSB above the last code transition (111 . . . 110 to

111 . . . 111). The error is expressed as a percentage of full

scale.

POSITIVE FULL-SCALE ERROR

Positive Full-Scale Error is the deviation of the last code transi-

tion (111 . . . 110 to 111 . . . 111) from the ideal AIN(+) voltage

(AIN(–) + V

REF

/GAIN – 3/2 LSBs). It applies to both unipolar

and bipolar analog input ranges.

UNIPOLAR OFFSET ERROR

Unipolar Offset Error is the deviation of the first code transition

from the ideal AIN(+) voltage (AIN(–) + 0.5 LSB) when oper-

ating in the unipolar mode.

BIPOLAR ZERO ERROR

This is the deviation of the midscale transition (0111 . . . 111

to 1000 . . . 000) from the ideal AIN(+) voltage (AIN(–) –

0.5 LSB) when operating in the bipolar mode.

GAIN ERROR

This is a measure of the span error of the ADC. It includes full-

scale errors but not zero-scale errors. For unipolar input ranges

it is defined as (full-scale error – unipolar offset error) while for

bipolar input ranges it is defined as (full-scale error – bipolar

zero error).

*AIN(–) refers to the negative input of the differential input pairs or to AIN6

when referring to the pseudo-differential input configurations.

AD7714

REV. C –11–

AD7714-5 OUTPUT NOISE

Table Ia shows the output rms noise and effective resolution for some typical notch and –3␣ dB frequencies for the AD7714-5 with

CLK␣ IN

= 2.4576␣ MHz while Table Ib gives the information for f

CLK IN

= 1␣ MHz. The numbers given are for the bipolar input ranges

with a V

REF

of +2.5␣ V and with BUFFER = 0. These numbers are typical and are generated at an analog input voltage of 0␣ V. The

numbers in brackets in each table are for the effective resolution of the part (rounded to the nearest 0.5␣ LSB). The effective resolu-

tion of the device is defined as the ratio of the output rms noise to the input full scale (i.e., 2 × V

REF

/GAIN). It should be noted that

it is not calculated using peak-to-peak output noise numbers. Peak-to-peak noise numbers can be up to 6.6 times the rms numbers

while effective resolution numbers based on peak-to-peak noise can be 2.5 bits below the effective resolution based on rms noise as

quoted in the tables.

The output noise from the part comes from two sources. The first is the electrical noise in the semiconductor devices used in the

implementation of the modulator (device noise). Secondly, when the analog input signal is converted into the digital domain, quan-

tization noise is added. The device noise is at a low level and is largely independent of frequency. The quantization noise starts at

an even lower level but rises rapidly with increasing frequency to become the dominant noise source. Consequently, lower filter

notch settings (below 100␣ Hz approximately for f

CLK IN

= 2.4576␣ MHz and below 40␣ Hz approximately for f

CLK IN

= 1␣ MHz) tend to

be device noise dominated while higher notch settings are dominated by quantization noise. Changing the filter notch and cutoff

frequency in the quantization-noise dominated region results in a more dramatic improvement in noise performance than it does in

the device-noise dominated region as shown in Table I. Furthermore, quantization noise is added after the PGA, so effective resolu-

tion is largely independent of gain for the higher filter notch frequencies. Meanwhile, device noise is added in the PGA and, there-

fore, effective resolution reduces at high gains for lower notch frequencies. Additionally, in the device-noise dominated region, the

output noise (in µV) is largely independent of reference voltage while in the quantization-noise dominated region, the noise is pro-

portional to the value of the reference. It is possible to do post-filtering on the device to improve the output data rate for a given

–3␣ dB frequency and also to further reduce the output noise.

At the lower filter notch settings (below 60␣ Hz for f

CLK IN

= 2.4576␣ MHz and below 25␣ Hz for f

CLK IN

= 1␣ MHz), the no missing

codes performance of the device is at the 24-bit level. At the higher settings, more codes will be missed until at 1␣ kHz notch setting

for f

CLK␣ IN

= 2.4576␣ MHz (400␣ Hz for f

CLK IN

= 1␣ MHz), no missing codes performance is only guaranteed to the 12-bit level.

Table Ia. AD7714-5 Output Noise/Resolution vs. Gain and First Notch for f

CLK IN

= 2.4576␣ MHz, BUFFER = 0

Filter First Typical Output RMS Noise in ␮V (Effective Resolution in Bits)

Notch & O/P –3␣ dB Gain of Gain of Gain of Gain of Gain of Gain of Gain of Gain of

Data Rate Frequency 1 2 4 8 16 32 64 128

5␣ Hz 1.31␣ Hz 0.87 (22.5) 0.48 (22.5) 0.24 (22.5) 0.2 (21.5) 0.18 (20.5) 0.17 (20) 0.17 (19) 0.17 (18)

10␣ Hz 2.62␣ Hz 1.0 (22.5) 0.78 (21.5) 0.48 (21.5) 0.33 (21) 0.25 (20.5) 0.25 (19.5) 0.25 (18.5) 0.25 (17.5)

25␣ Hz 6.55␣ Hz 1.8 (21.5) 1.1 (21) 0.63 (21) 0.5 (20) 0.44 (19.5) 0.41 (18.5) 0.38 (17.5) 0.38 (16.5)

30␣ Hz 7.86␣ Hz 2.5 (21) 1.31 (21) 0.84 (20.5) 0.57 (20) 0.46 (19.5) 0.43 (18.5) 0.4 (17.5) 0.4 (16.5)

50␣ Hz 13.1␣ Hz 4.33 (20) 2.06 (20) 1.2 (20) 0.64 (20) 0.54 (19) 0.46 (18.5) 0.46 (17.5) 0.46 (16.5)

60␣ Hz 15.72␣ Hz 5.28 (20) 2.36 (20) 1.33 (20) 0.87 (19.5) 0.63 (19) 0.62 (18) 0.6 (17) 0.56 (16)

100␣ Hz 26.2␣ Hz 12.1 (18.5) 5.9 (18.5) 2.86 (19) 1.91 (18.5) 1.06 (18) 0.83 (17.5) 0.82 (16.5) 0.76 (15.5)

250␣ Hz 65.5␣ Hz 127 (15.5) 58 (15.5) 29 (15.5) 15.9 (15.5) 6.7 (15.5) 3.72 (15.5) 1.96 (15.5) 1.5 (14.5)

500␣ Hz 131␣ Hz 533 (13) 267 (13) 137 (13) 66 (13) 38 (13) 20 (13) 8.6 (13) 4.4 (13)

1␣ kHz 262␣ Hz 2,850 (11) 1,258 (11) 680 (11) 297 (11) 131 (11) 99 (10.5) 53 (10.5) 28 (10.5)

Table Ib. AD7714-5 Output Noise/Resolution vs. Gain and First Notch for f

CLK IN

= 1␣ MHz, BUFFER = 0

Filter First Typical Output RMS Noise in ␮V (Effective Resolution in Bits)

Notch & O/P –3␣ dB Gain of Gain of Gain of Gain of Gain of Gain of Gain of Gain of

Data Rate Frequency 1 2 4 8 16 32 64 128

2␣ Hz 0.52␣ Hz 0.75 (22.5) 0.56 (22) 0.31 (22) 0.19 (21.5) 0.17 (21) 0.14 (20) 0.14 (19) 0.14 (18)

4␣ Hz 1.05␣ Hz 1.04 (22) 0.88 (21.5) 0.45 (21.5) 0.28 (21) 0.21 (20.5) 0.21 (19.5) 0.21 (18.5) 0.21 (17.5)

10␣ Hz 2.62␣ Hz 1.66 (21.5) 1.01 (21.5) 0.77 (20.5) 0.41 (20.5) 0.37 (19.5) 0.35 (19) 0.35 (18) 0.35 (17)

25 Hz 6.55␣ Hz 5.2 (20) 2.06 (20) 1.4 (20) 0.86 (19.5) 0.63 (19) 0.61 (18) 0.59 (17) 0.59 (16)

30␣ Hz 7.86␣ Hz 7.1 (19.5) 3.28 (19.5) 1.42 (19.5) 1.07 (19) 0.78 (18.5) 0.64 (18) 0.61 (17) 0.61 (16)

50␣ Hz 13.1␣ Hz 19.4 (18) 9.11 (18) 4.2 (18) 2.45 (18) 1.56 (17.5) 1.1 (17) 0.82 (16.5) 0.8 (15.5)

60␣ Hz 15.72␣ Hz 25 (17.5) 16 (17.5) 6.5 (17.5) 2.9 (17.5) 1.93 (17.5) 1.4 (17) 1.1 (16) 0.98 (15.5)

100␣ Hz 26.2␣ Hz 102 (15.5) 58 (15.5) 25 (15.5) 13.5 (15.5) 5.7 (15.5) 3.9 (15.5) 2.1 (15) 1.3 (15)

200␣ Hz 52.4␣ Hz 637 (13) 259 (13) 130 (13) 76 (13) 33 (13) 16 (13) 11 (13) 6 (12.5)

400␣ Hz 104.8␣ Hz 2,830 (11) 1,430 (11) 720 (11) 334 (11) 220 (10.5) 94 (10.5) 54 (10.5) 25 (10.5)

AD7714

REV. C–12–

AD7714-3 OUTPUT NOISE

Table IIa shows the output rms noise and effective resolution for some typical notch and –3␣ dB frequencies for the AD7714-3 with

CLK␣ IN

= 2.4576␣ MHz while Table IIb gives the information for f

CLK IN

= 1␣ MHz. The numbers given are for the bipolar input

ranges with a V

REF

of +1.25␣ V and BUFFER = 0. These numbers are typical and are generated at an analog input voltage of 0␣ V.

The numbers in brackets in each table are for the effective resolution of the part (rounded to the nearest 0.5␣ LSB). The effective

resolution of the device is defined as the ratio of the output rms noise to the input full scale (i.e., 2 × V

REF

/GAIN). It should be

noted that it is not calculated using peak-to-peak output noise numbers. Peak-to-peak noise numbers can be up to 6.6 times the rms

numbers while effective resolution numbers based on peak-to-peak noise can be 2.5 bits below the effective resolution based on rms

noise as quoted in the tables.

The output noise from the part comes from two sources. The first is the electrical noise in the semiconductor devices used in the

implementation of the modulator (device noise). Secondly, when the analog input signal is converted into the digital domain, quan-

tization noise is added. The device noise is at a low level and is largely independent of frequency. The quantization noise starts at

an even lower level but rises rapidly with increasing frequency to become the dominant noise source. Consequently, lower filter

notch settings (below 100␣ Hz approximately for f

CLK IN

= 2.4576␣ MHz and below 40␣ Hz approximately for f

CLK IN

= 1␣ MHz) tend to

be device noise dominated while higher notch settings are dominated by quantization noise. Changing the filter notch and cutoff

frequency in the quantization noise dominated region results in a more dramatic improvement in noise performance than it does in

the device-noise dominated region as shown in Table II. Furthermore, quantization noise is added after the PGA, so effective reso-

lution is largely independent of gain for the higher filter notch frequencies. Meanwhile, device noise is added in the PGA and, there-

fore, effective resolution suffers a little at high gains for lower notch frequencies. Additionally, in the device-noise dominated region,

the output noise (in µV) is largely independent of reference voltage while in the quantization-noise dominated region, the noise is

proportional to the value of the reference. It is possible to do post-filtering on the device to improve the output data rate for a given

–3␣ dB frequency and also to further reduce the output noise.

At the lower filter notch settings (below 60␣ Hz for f

CLK IN

= 2.4576␣ MHz and below 25␣ Hz for f

CLK IN

= 1␣ MHz), the no missing

codes performance of the device is at the 24-bit level. At the higher settings, more codes will be missed until at 1␣ kHz notch setting

for f

CLK␣ IN

= 2.4576␣ MHz (400␣ Hz for f

CLK IN

= 1␣ MHz), no missing codes performance is only guaranteed to the 12-bit level.

Filter First Typical Output RMS Noise in ␮V (Effective Resolution in Bits)

Notch & O/P –3␣ dB Gain of Gain of Gain of Gain of Gain of Gain of Gain of Gain of

Data Rate Frequency 1 2 4 8 16 32 64 128

2␣ Hz 0.52␣ Hz 0.86 (21.5) 0.58 (21) 0.32 (21) 0.21 (20.5) 0.2 (19.5) 0.2 (18.5) 0.2 (17.5) 0.2 (16.5)

4␣ Hz 1.05␣ Hz 1.26 (21) 0.74 (20.5) 0.44 (20.5) 0.35 (20) 0.3 (19) 0.3 (18) 0.3 (17) 0.3 (16)

10␣ Hz 2.62␣ Hz 1.68 (20.5) 1.33 (20) 0.73 (20) 0.5 (19) 0.49 (18.5) 0.49 (17.5) 0.48 (16.5) 0.47 (15.5)

25␣ Hz 6.55␣ Hz 3.82 (19.5) 2.0 (19.5) 1.2 (19) 0.88 (18.5) 0.66 (18) 0.57 (17) 0.55 (16) 0.55 (15)

30␣ Hz 7.86␣ Hz 4.88 (19) 2.1 (19) 1.3 (19) 0.93 (18.5) 0.82 (17.5) 0.69 (17) 0.68 (16) 0.66 (15)

50␣ Hz 13.1␣ Hz 11 (18) 4.8 (18) 2.4 (18) 1.4 (18) 1.4 (17) 0.73 (16.5) 0.71 (15.5) 0.7 (15)

60␣ Hz 15.72␣ Hz 14.7 (17.5) 7.5 (17.5) 3.8 (17.5) 2.6 (17) 1.5 (16.5) 0.95 (16.5) 0.88 (15) 0.9 (14.5)

100␣ Hz 26.2␣ Hz 61 (15.5) 30 (15.5) 12 (15.5) 6.1 (15.5) 2.9 (15.5) 2.4 (15) 1.8 (14.5) 1.8 (13.5)

200␣ Hz 52.4␣ Hz 275 (13) 130 (13) 65 (13) 33 (13) 17 (13) 11 (13) 6.3 (12.5) 3 (12.5)

400 Hz 104.8␣ Hz 1435 (11) 720 (11) 362 (11) 175 (11) 110 (10.5) 51 (10.5) 31 (10.5) 12 (10.5)

Table IIa. AD7714-3 Output Noise/Resolution vs. Gain and First Notch for f

CLK IN

= 2.4576␣ MHz, BUFFER = 0

Filter First Typical Output RMS Noise in ␮V (Effective Resolution in Bits)

Notch & O/P –3␣ dB Gain of Gain of Gain of Gain of Gain of Gain of Gain of Gain of

Data Rate Frequency 1 2 4 8 16 32 64 128

5␣ Hz 1.31␣ Hz 1.07 (21) 0.68 (21) 0.29 (21) 0.24 (20) 0.22 (19.5) 0.22 (18.5) 0.22 (17.5) 0.22 (16.5)

10␣ Hz 2.62␣ Hz 1.69 (20.5) 1.1 (20) 0.56 (20) 0.35 (19.5) 0.33 (19) 0.33 (18) 0.33 (17) 0.33 (16)

25␣ Hz 6.55␣ Hz 3.03 (19.5) 1.7 (19.5) 0.89 (19.5) 0.55 (19) 0.49 (18.5) 0.46 (17.5) 0.46 (16.5) 0.45 (15.5)

30␣ Hz 7.86␣ Hz 3.55 (19.5) 2.1 (19) 1.1 (19) 0.61 (18.5) 0.58 (18) 0.57 (17) 0.55 (16) 0.55 (15)

50␣ Hz 13.1␣ Hz 4.72 (19) 2.3 (19) 1.5 (18.5) 0.84 (18.5) 0.7 (18) 0.68 (17) 0.67 (16) 0.66 (15)

60␣ Hz 15.72␣ Hz 5.12 (19) 3.1 (18.5) 1.6 (18) 0.98 (18) 0.9 (17.5) 0.7 (17) 0.69 (16) 0.68 (15)

100␣ Hz 26.2␣ Hz 9.68 (18) 5.6 (18) 2.4 (18) 1.3 (18) 1.1 (17) 0.95 (16.5) 0.88 (15.5) 0.9 (14.5)

250␣ Hz 65.5␣ Hz 44 (16) 31 (15.5) 15 (15.5) 5.8 (15.5) 3.7 (15.5) 2.4 (15) 1.8 (14.5) 1.8 (13.5)

500␣ Hz 131␣ Hz 304 (13) 129 (13) 76 (13) 33 (13) 20 (13) 11 (13) 6.3 (12.5) 3 (12.5)

1␣ kHz 262␣ Hz 1410 (11) 715 (11) 350 (11) 177 (11) 101 (10.5) 51 (10.5) 31 (10.5) 12 (10.5)

Table IIb. AD7714-3 Output Noise/Resolution vs. Gain and First Notch for f

CLK IN

= 1␣ MHz, BUFFER = 0

AD7714

REV. C –13–

BUFFERED MODE NOISE

Table III shows the typical output rms noise and effective resolution for some typical notch and –3␣ dB frequencies for the AD7714-

5 with f

CLK␣ IN

= 2.4576␣ MHz and BUFFER = +5 V. Table IV gives the information for the AD7714-3 again with f

CLK IN

= 2.4576

MHz and BUFFER = +5␣ V. The numbers given are for the bipolar input ranges and are generated with a differential analog input

voltage of 0␣ V. For the AD7714-5, the V

REF

voltage is +2.5␣ V while for the AD7714 the V

REF

voltage is +1.25␣ V. The numbers in

brackets in each table are for the effective resolution of the part (rounded to the nearest 0.5 LSB). The effective resolution of the

device is defined as the ratio of the output rms noise to the input full scale (i.e., 2 × V

REF

/GAIN). It should be noted that it is not

calculated using peak-to-peak output noise numbers. Peak-to-peak noise numbers can be up to 6.6 times the rms numbers while

effective resolution numbers based on peak-to-peak noise can be 2.5 bits below the effective resolution based on rms noise as quoted

in the tables.

Table III. AD7714-5 Buffered Mode Output Noise/Resolution for f

CLK IN

= 2.4576␣ MHz

Filter First Typical Output RMS Noise in ␮V (Effective Resolution in Bits)

Notch & O/P –3␣ dB Gain of Gain of Gain of Gain of Gain of Gain of Gain of Gain of

Data Rate Frequency 1 2 4 8 16 32 64 128

5␣ Hz 1.31␣ Hz 0.99 (22.5) 0.68 (22) 0.46 (21.5) 0.26 (21) 0.26 (20) 0.26 (19) 0.26 (18) 0.26 (17)

10␣ Hz 2.62␣ Hz 1.5 (21.5) 0.95 (21.5) 0.63 (21) 0.41 (20.5) 0.39 (19.5) 0.36 (18.5) 0.36 (17.5) 0.36 (16.5)

25␣ Hz 6.55␣ Hz 2.5 (21) 1.7 (20.5) 0.88 (20.5) 0.75 (19.5) 0.57 (19) 0.57 (18) 0.57 (17) 0.56 (16)

30␣ Hz 7.86␣ Hz 2.9 (20.5) 1.8 (20.5) 1 (20) 0.87 (19.5) 0.75 (18.5) 0.72 (17.5) 0.72 (16.5) 0.71 (15.5)

50␣ Hz 13.1␣ Hz 4.2 (20) 2.5 (20) 1.5 (19.5) 1.1 (19) 0.94 (18.5) 0.94 (17.5) 0.94 (16.5) 0.87 (15.5)

60␣ Hz 15.72␣ Hz 6.1 (19.5) 2.9 (19.5) 2 (19.5) 1.2 (19) 1 (18.5) 0.97 (17.5) 0.95 (16.5) 0.94 (15.5)

100␣ Hz 26.2␣ Hz 13.8 (18.5) 6.5 (18.5) 3.5 (18.5) 2.2 (18) 1.3 (18) 1.2 (17) 1.3 (16) 1.1 (15)

250␣ Hz 65.5␣ Hz 87 (16) 56 (15.5) 25 (15.5) 11 (15.5) 5.7 (15.5) 3.6 (15.5) 2.4 (15) 2.1 (14)

500␣ Hz 131␣ Hz 508 (13.5) 241 (13.5) 117 (13.5) 73 (13) 34 (13) 16 (13) 8.5 (13) 5.2 (13)

1␣ kHz 262␣ Hz 2860 (11) 1700 (10.5) 745 (10.5) 480 (10.5) 197 (10.5) 94 (10.5) 53 (10.5) 23 (10.5)

Table IV. AD7714-3 Buffered Mode Output Noise/Resolution for f

CLK IN

= 2.4576␣ MHz

Filter First Typical Output RMS Noise in ␮V (Effective Resolution in Bits)

Notch & O/P –3␣ dB Gain of Gain of Gain of Gain of Gain of Gain of Gain of Gain of

Data Rate Frequency 1 2 4 8 16 32 64 128

5␣ Hz 1.31␣ Hz 1.16 (21) 0.76 (20.5) 0.34 (20) 0.29 (20) 0.29 (19) 0.28 (18) 0.26 (17) 0.26 (16)

10␣ Hz 2.62␣ Hz 1.7 (20.5) 1 (20.5) 0.7 (20) 0.46 (19.5) 0.45 (18.5) 0.4 (17.5) 0.4 (16.5) 0.4 (15.5)

25␣ Hz 6.55␣ Hz 3.5 (19.5) 1.8 (19.5) 1.1 (19) 0.74 (18.5) 0.63 (18) 0.6 (17) 0.6 (16) 0.6 (15)

30␣ Hz 7.86␣ Hz 3.7 (19.5) 2.2 (19) 1.3 (19) 0.76 (18.5) 0.68 (18) 0.66 (17) 0.66 (16) 0.66 (15)

50␣ Hz 13.1␣ Hz 4.5 (19) 3 (18.5) 1.7 (18.5) 1.0 (18) 0.92 (17.5) 0.9 (16.5) 0.89 (15.5) 0.89 (14.5)

60␣ Hz 15.72␣ Hz 5.3 (19) 3.3 (18.5) 1.8 (18.5) 1.1 (18) 1 (17) 0.96 (16.5) 0.96 (15.5) 0.96 (14.5)

100␣ Hz 26.2␣ Hz 10 (18) 4.9 (18) 3.1 (17.5) 1.5 (17.5) 1.2 (17) 1.2 (16) 1.2 (15) 1.2 (14)

250␣ Hz 65.5␣ Hz 47 (15.5) 29 (15.5) 15 (15.5) 7.5 (15.5) 4.7 (15) 2.6 (15) 2.5 (14) 1.6 (13.5)

500␣ Hz 131␣ Hz 300 (13.5) 171 (13) 74 (13) 35 (13) 21 (13) 8.6 (13) 5.6 (13) 3.1 (12.5)

1␣ kHz 262␣ Hz 1722 (10.5) 735 (10.5) 380 (10.5) 230 (10.5) 93 (10.5) 55 (10.5) 30 (10.5) 12 (10.5)

AD7714

REV. C–14–

ON-CHIP REGISTERS

The AD7714 contains eight on-chip registers which can be accessed via the serial port of the part. The first of these is a Communica-

tions Register which controls the channel selection, decides whether the next operation is a read or write operation and also decides

which register the next read or write operation accesses. All communications to the part must start with a write operation to the

Communications Register. After power-on or RESET, the device expects a write to its Communications Register. The data written

to this register determines whether the next operation to the part is a read or a write operation and also determines to which register

this read or write operation occurs. Therefore, write access to any of the other registers on the part starts with a write operation to the

Communications Register followed by a write to the selected register. A read operation from any other register on the part (including

the output data register) starts with a write operation to the Communications Register followed by a read operation from the selected

register. The communications register also controls channel selection and the DRDY status is also available by reading from the

Communications Register. The second register is a Mode Register which determines calibration mode and gain setting. The third

register is labelled the Filter High Register and this determines the word length, bipolar/unipolar operation and contains the upper 4

bits of the filter selection word. The fourth register is labelled the Filter Low Register and contains the lower 8 bits of the filter selec-

tion word. The fifth register is a Test Register which is accessed when testing the device. The sixth register is the Data Register from

which the output data from the part is accessed. The final registers allow access to the part’s calibration registers. The Zero Scale

Calibration Register allows access to the zero scale calibration coefficients for the selected input channel while the Full Scale Calibra-

tion Register allows access to the full scale calibration coefficients for the selected input channel. The registers are discussed in more

detail in the following sections.

Communications Register (RS2-RS0 = 0, 0, 0)

The Communications Register is an 8-bit register from which data can either be read or to which data can be written. All communi-

cations to the part must start with a write operation to the Communications Register. The data written to the Communications Reg-

ister determines whether the next operation is a read or write operation and to which register this operation takes place. Once the

subsequent read or write operation to the selected register is complete, the interface returns to where it expects a write operation to

the Communications Register. This is the default state of the interface, and on power-up or after a RESET, the AD7714 is in this

default state waiting for a write operation to the Communications Register. In situations where the interface sequence is lost, if a

write operation of sufficient duration (containing at least 32 serial clock cycles) takes place with DIN high, the AD7714 returns to

this default state. Table V outlines the bit designations for the Communications Register.

Table V. Communications Register

Table VI. Register Selection

0/DRDY RS2 RS1 RS0 R/WCH2 CH1 CH0

0/DRDY For a write operation, a 0 must be written to this bit so that the write operation to the Communications Register

actually takes place. If a 1 is written to this bit, the part will not clock on to subsequent bits in the register. It will stay

at this bit location until a 0 is written to this bit. Once a 0 is written to this bit, the next 7 bits will be loaded to the

Communications Register. For a read operation, this bit provides the status of the DRDY flag from the part. The

status of this bit is the same as the DRDY output pin.

RS2–RS0 Register Selection Bits. RS2 is the MSB of the three selection bits. The three bits select to which one of eight on-chip

registers the next read or write operation takes place as shown in Table VI along with the register size.

RS2 RS1 RS0 Register Register Size

0 0 0 Communications Register 8 Bits

0 0 1 Mode Register 8 Bits

0 1 0 Filter High Register 8 Bits

0 1 1 Filter Low Register 8 Bits

1 0 0 Test Register 8 Bits

1 0 1 Data Register 16 Bits or 24 Bits

1 1 0 Zero-Scale Calibration Register 24 Bits

1 1 1 Full-Scale Calibration Register 24 Bits

AD7714

REV. C –15–

CH2 CH1 CH0 AIN(+) AIN(–) Type Calibration Register Pair

0 0 0 AIN1 AIN6 Pseudo Differential Register Pair 0

0 0 1 AIN2 AIN6 Pseudo Differential Register Pair 1

0 1 0 AIN3 AIN6 Pseudo Differential Register Pair 2

0 1 1 AIN4 AIN6 Pseudo Differential Register Pair 2

1 0 0 AIN1 AIN2 Fully Differential Register Pair 0

1 0 1 AIN3 AIN4 Fully Differential Register Pair 1

1 1 0 AIN5 AIN6 Fully Differential Register Pair 2

1 1 1 AIN6 AIN6 Test Mode Register Pair 2

Mode Register (RS2-RS0 = 0, 0, 1); Power On/Reset Status: 00␣ Hex

The Mode Register is an eight bit register from which data can either be read or to which data can be written. Table VIII outlines the

bit designations for the Mode Register.

Table VIII. Mode Register

MD2 MD1 MD0 G2 G1 G0 BO FSYNC

MD2 MD1 MD0 Operating Mode

0 0 0 Normal Mode; this is the normal mode of operation of the device whereby the device is performing nor-

mal conversions. This is the default condition of these bits after Power-On or RESET.

0 0 1 Self-Calibration; this activates self-calibration on the channel selected by CH2, CH1 and CH0 of the

Communications Register. This is a one step calibration sequence and when complete the part returns to

Normal Mode with MD2, MD1 and MD0 returning to 0, 0, 0. The DRDY output or bit goes high when

calibration is initiated and returns low when this self-calibration is complete and a new valid word is

available in the data register. The zero-scale calibration is performed at the selected gain on internally

shorted (zeroed) inputs and the full-scale calibration is performed at the selected gain on an internally-

generated V

REF

/Selected Gain.

0 1 0 Zero-Scale System Calibration; this activates zero scale system calibration on the channel selected by

CH2, CH1 and CH0 of the Communications Register. Calibration is performed at the selected gain on

the input voltage provided at the analog input during this calibration sequence. This input voltage should

remain stable for the duration of the calibration. The DRDY output or bit goes high when calibration is

initiated and returns low when this zero-scale calibration is complete and a new valid word is available in

the data register. At the end of the calibration, the part returns to Normal Mode with MD2, MD1 and

MD0 returning to 0, 0, 0.

0 1 1 Full-Scale System Calibration; this activates full-scale system calibration on the selected input channel.

Calibration is performed at the selected gain on the input voltage provided at the analog input during this

calibration sequence. This input voltage should remain stable for the duration of the calibration. Once

again, the DRDY output or bit goes high when calibration is initiated and returns low when this full-scale

calibration is complete and a new valid word is available in the data register. At the end of the calibration,

the part returns to Normal Mode with MD2, MD1 and MD0 returning to 0, 0, 0.

Table VII. Channel Selection

CH2–CH0 Channel Select. These three bits select a channel either for conversion or for access to calibration coefficients as

outlined in Table VII. There are three pairs of calibration registers on the part. In fully differential mode, the part

has three input channels so each channel has its own pair of calibration registers. In pseudo-differential mode, the

AD7714 has five input channels with some of the input channel combinations sharing calibration registers. With

CH2, CH1 and CH0 at a logic 1, the part looks at the AIN6 input internally shorted to itself. This can be used as

a test method to evaluate the noise performance of the part with no external noise sources. In this mode, the AIN6

input should be connected to an external voltage within the allowable common-mode range for the part. The

Power-On or RESET status of these bits is 1,0,0 selecting the differential pair AIN1 and AIN2.

AD7714

REV. C–16–

MD2 MD1 MD0 Operating Mode (continued)

1 0 0 System-Offset Calibration; this activates system-offset calibration on the channel selected by CH2, CH1

and CH0 of the Communications Register. This is a one step calibration sequence and when complete

the part returns to Normal Mode with MD2, MD1 and MD0 returning to 0, 0, 0. The DRDY output

or bit goes high when calibration is initiated and returns low when this system offset calibration is com-

plete and a new valid word is available in the data register. For this calibration type, the zero-scale cali-

bration is performed at the selected gain on the input voltage provided at the analog input during this

calibration sequence. This input voltage should remain stable for the duration of the calibration. The

full-scale calibration is performed at the selected gain on an internally generated V

REF

/Selected Gain.

1 0 1 Background Calibration; this activates background calibration on the channel selected by CH2, CH1

and CH0 of the Communications Register. If the background calibration mode is on, then the AD7714

provides continuous self-calibration of the shorted (zeroed) inputs. This calibration takes place as part

of the conversion sequence, extending the conversion time and reducing the word rate by a factor of six.

Its major advantage is that the user does not have to worry about recalibrating the offset of the device

when there is a change in the ambient temperature or supplies. In this mode, the zero-scale calibration

is performed at the selected gain on internally shorted (zeroed) inputs. The calibrations are interleaved

with normal conversions and the calibration registers of the device are automatically updated. Because

the background calibration does not perform full-scale calibrations, a self-calibration should be per-

formed before placing the part in the background calibration mode.

1 1 0 Zero-Scale Self-Calibration; this activates zero-scale self-calibration on the channel selected by CH2,

CH1 and CH0 of the Communications Register. This zero-scale self-calibration is performed at the

selected gain on internally shorted (zeroed) inputs. This is a one step calibration sequence and when

complete the part returns to Normal Mode with MD2, MD1 and MD0 returning to 0, 0, 0. The DRDY

output or bit goes high when calibration is initiated and returns low when this zero-scale self-calibration

is complete and a new valid word is available in the data register.

1 1 1 Full-Scale Self-Calibration; this activates full-scale self-calibration on the channel selected by CH2,

CH1 and CH0 of the Communications Register. This full-scale self-calibration is performed at the

selected gain on an internally-generated V

REF

/Selected Gain. This is a one step calibration sequence and

when complete the part returns to Normal Mode with MD2, MD1 and MD0 returning to 0, 0, 0. The

DRDY output or bit goes high when calibration is initiated and returns low when this full-scale self-

calibration is complete and a new valid word is available in the data register.

G2 G1 G0 Gain Setting

0001

0012

0104

0118

10016

10132

11064

111128

BO Burnout Current. A 0 in this bit turns off the on-chip burnout currents. This is the default (Power-On

or RESET) status of this bit. A 1 in this bit activates the burnout currents. When active, the burnout

currents connect to the selected analog input pair, one to the AIN(+) input and one to the AIN(–) input.

FSYNC Filter Synchronization. When this bit is high, the nodes of the digital filter, the filter control logic and

the calibration control logic are held in a reset state and the analog modulator is also held in its reset

state. When this bit goes low, the modulator and filter start to process data and a valid word is available

in 3 × 1/(output update rate), i.e., the settling time of the filter. This FSYNC bit does not affect the

digital interface and does not reset the DRDY output if it is low.

AD7714

REV. C –17–

Filter Registers. Power On/Reset Status: Filter High Register: 01␣ Hex. Filter Low Register: 40␣ Hex.

There are two 8-bit Filter Registers on the AD7714 from which data can either be read or to which data can be written. Tables IX

and X outline the bit designations for the Filter Registers.

Table IX. Filter High Register (RS2–RS0 = 0, 1, 0)

B/U WL BST ZERO FS11 FS10 FS9 FS8 A Versions

B/U WL BST CLKDIS FS11 FS10 FS9 FS8 Y Versions

Table X. Filter Low Register (RS2–RS0 = 0, 1, 1)

FS7 FS6 FS5 FS4 FS3 FS2 FS1 FS0 All Versions

B/U Bipolar/Unipolar Operation. A 0 in this bit selects Bipolar Operation. This is the default (Power-On or RESET)

status of this bit. A 1 in this bit selects unipolar operation.

WL Word Length. A 0 in this bit selects 16-bit word length when reading from the data register (i.e., DRDY returns

high after 16 serial clock cycles in the read operation). This is the default (Power-On or RESET) status of this

bit. A 1 in this bit selects 24-bit word length.

BST Current Boost. A 0 in this bit reduces the current taken by the analog front end. When the part is operated with

CLK IN

= 1␣ MHz or at gains of 1 to 4 with f

CLK IN

= 2.4576␣ MHz, this bit should be 0 to reduce the current

drawn from AV

, although the device will operate just as well with this bit at a 1. When the AD7714 is oper-

ated at gains of 8 to 128 with f

CLK IN

= 2.4576␣ MHz, this bit must be 1 to ensure correct operation of the

device. The Power-On or RESET status of this bit is 0.

ZERO To ensure correct operation of the A Versions of the part, a 0 must be written to this bit.

CLKDIS Master Clock Disable Bit. A Logic 1 in this bit disables the master clock from appearing at the MCLKOUT

pin. When disabled, the MCLKOUT pin is forced low. This feature allows the user the flexibility of using the

MCLKOUT as a clock source for other devices in the system or for turning off the MCLKOUT as a power

saving feature. When using an external master clock or the MCLKIN pin, the AD7714 continues to have inter-

nal clocks and will convert normally with its CLKDIS bit active. When using a crystal oscillator or ceramic

resonator across the MCLK IN or MCLKOUT pins, the AD7714 clock is stopped and no conversions take

place when the CLKDIS bit is active.

FS11–FS0 Filter Selection. The on-chip digital filter provides a Sinc

(or (Sinx/x)

) filter response. The 12 bits of data

programmed into these bits determine the filter cut-off frequency, the position of the first notch of the filter and

the data rate for the part. In association with the gain selection, it also determines the output noise (and hence

the effective resolution) of the device.

The first notch of the filter occurs at a frequency determined by the relationship:

filter first notch frequency =␣ (f

CLK␣ IN

/128)/code

where code is the decimal equivalent of the code in bits FS0 to FS11 and is in the range 19 to 4,000. With the

nominal f

CLK IN

of 2.4576␣ MHz, this results in a first notch frequency range from 4.8␣ Hz to 1.01␣ kHz. To

ensure correct operation of the AD7714, the value of the code loaded to these bits must be within this range.

Failure to do this will result in unspecified operation of the device.

Changing the filter notch frequency, as well as the selected gain, impacts resolution. Tables I through IV show

the effect of the filter notch frequency and gain on the effective resolution of the AD7714. The output data rate

(or effective conversion time) for the device is equal to the frequency selected for the first notch of the filter. For

example, if the first notch of the filter is selected at 50␣ Hz then a new word is available at a 50 Hz rate or every

20␣ ms. If the first notch is at 1␣ kHz, a new word is available every 1␣ ms.

The settling time of the filter to a full-scale step input change is worst case 4 × 1/(output data rate). For

example, with the first filter notch at 50␣ Hz, the settling time of the filter to a full-scale step input change is

80␣ ms max. This settling time can be reduced to 3 × 1/(output data rate) by synchronizing the step input

change to a reset of the digital filter. In other words, if the step input takes place with the SYNC input low or

the FSYNC bit high, the settling time will be 3 × 1/(output data rate) from when SYNC returns high or

FSYNC returns low. If a change of channel takes place, the settling time is 3 × 1/(output data rate) regardless of

the SYNC or FSYNC status as the part issues an internal SYNC command when requested to change channels.

The –3 dB frequency is determined by the programmed first notch frequency according to the relationship:

filter –3 dB frequency = 0.262 × filter first notch frequency.

AD7714

REV. C–18–

Test Register (RS2–RS0 = 1, 0, 0)

The part contains a Test Register which is used in testing the device. The user is advised not to change the status of any of the bits in this

register from the default (Power-On or RESET) status of all 0s as the part will be placed in one of its test modes and will not operate cor-

rectly. If the part enters one of its test modes, exercising RESET will exit the part from the mode. An alternative scheme for getting the

part out of one of its test modes, is to reset the interface by writing 32 successive 1s to the part and then write all 0s to the Test Register.

Data Register (RS2–RS0 = 1, 0, 1)

The Data Register on the part is a read-only register which contains the most up-to-date conversion result from the AD7714. The

register can be programmed to be either 16-bits or 24-bits wide, determined by the status of the WL bit of the Mode Register. If the

Communications Register data sets up the part for a write operation to this register, a write operation must actually take place in

order to return the part to where it is expecting a write operation to the Communications Register (the default state of the interface).

However, the 16 or 24 bits of data written to the part will be ignored by the AD7714.

Zero-Scale Calibration Register (RS2–RS0 = 1, 1, 0); Power On/Reset Status: 1F4000␣ Hex

The AD7714 contains three zero-scale calibration registers, labelled Zero-Scale Calibration Register 0 to Zero Scale Calibration

Register␣ 2. The three registers are totally independent of each other such that in fully differential mode there is a zero-scale register

for each of the input channels. Each of these registers is a 24-bit read/write register and, when writing to the registers, 24 bits must be

written; otherwise no data will be transferred to the register. The register is used in conjunction with the associated full-scale calibra-

tion register to form a register pair. These register pairs are associated with input channel pairs as outlined in Table VII.

While the part is set up to allow access to these registers over the digital interface, the part itself no longer has access to the register

coefficients to correctly scale the output data. As a result, there is a possibility that after accessing the calibration registers (either read

or write operation) the first output data read from the part may contain incorrect data. In addition, a read or write operation to the

calibration register should not be attempted while a calibration is in progress. These eventualities can be avoided by taking either the

SYNC input low or the FSYNC bit of the Mode Register high before the calibration register operation and taking them either high or

low respectively after the operation is complete.

Full-Scale Calibration Register (RS2–RS0 = 1, 1, 1); Power On/Reset Status: 5761AB␣ Hex

The AD7714 contains three full-scale calibration registers, labelled Full-Scale Calibration Register 0 to Full-Scale Calibration Regis-

ter 2. The three registers are totally independent of each other such that in fully differential mode there is a full-scale register for each

of the input channels. Each of these registers is a 24-bit read/write register and, when writing to the registers, 24 bits must be written,

otherwise no data will be transferred to the register. The register is used in conjunction with the associated zero-scale calibration

register to form a register pair. These register pairs are associated with input channel pairs as outlined in Table␣ VII.

While the part is set up to allow access to these registers over the digital interface, the part itself no longer has access to the coeffi-

cients to correctly scale the output data. As a result, there is a possibility that after accessing the calibration registers (either read or

write operation) the first output data read from the part may contain incorrect data. In addition, a read or write operation to the

calibration register should not be attempted while a calibration is in progress. These eventualities can be avoided by taking either the

SYNC input low or the FSYNC bit of the Mode Register high before the calibration register operation and taking them either high or

low respectively after the operation is complete.

CALIBRATION OPERATIONS

The AD7714 contains a number of calibration options as outlined previously. Table XI summarizes the calibration types, the opera-

tions involved and the duration of the operations. There are two methods of determining the end of calibration. The first is to moni-

tor when DRDY returns low at the end of the sequence. DRDY not only indicates when the sequence is complete but also that the

part has a valid new sample in its data register. This valid new sample is the result of a normal conversion which follows the calibra-

tion sequence. The second method of determining when calibration is complete is to monitor the MD2, MD1 and MD0 bits of the

Mode Register. When these bits return to 0, 0, 0 following a calibration command, it indicates that the calibration sequence is com-

plete. This method does not give any indication of there being a valid new result in the data register. However, it gives an earlier

indication that calibration is complete than DRDY. The time to when the Mode Bits (MD2, MD1 and MD0) return to 0, 0, 0

represents the duration of the calibration. The sequence to when DRDY goes low also includes a normal conversion and a pipeline

delay, t

(2000 × t

CLK IN

), to correctly scale the results of this first conversion. The time for both methods is given in the table.

Table XI. Calibration Operations

Calibration Type MD2, MD1, MD0 Calibration Sequence Duration to Mode Bits Duration to DRDY

Self Calibration 0, 0, 1 Internal ZS Cal @ Selected Gain + 6 × 1/Output Rate 9 × 1/Output Rate + t

Internal FS Cal @ Selected Gain

ZS System Calibration 0, 1, 0 ZS Cal on AIN @ Selected Gain 3 × 1/Output Rate 4 × 1/Output Rate + t

FS System Calibration 0, 1, 1 FS Cal on AIN @ Selected Gain 3 × 1/Output Rate 4 × 1/Output Rate + t

System-Offset Calibration 1, 0, 0 ZS Cal on AIN @ Selected Gain + 6 × 1/Output Rate 9 × 1/Output Rate + t

Internal FS Cal @ Selected Gain

Background Calibration 1, 0, 1 Internal ZS Cal @ Selected Gain + Bits Not Reset 6 × 1/Output Rate

Normal Conversion

ZS Self Calibration 1, 1, 0 Internal ZS Cal @ Selected Gain 3 × 1/Output Rate 6 × 1/Output Rate + t

FS Self Calibration 1, 1, 1 Internal FS Cal @ Selected Gain 3 × 1/Output Rate 6 × 1/Output Rate + t

AD7714

REV. C –19–

CIRCUIT DESCRIPTION

The AD7714 is a sigma-delta A/D converter with on-chip digi-

tal filtering, intended for the measurement of wide dynamic

range, low frequency signals such as those in weigh-scale, pres-

sure transducer, industrial control or process control applica-

tions. It contains a sigma-delta (or charge-balancing) ADC, a

calibration microcontroller with on-chip static RAM, a clock

oscillator, a digital filter and a bidirectional serial communica-

tions port. The part consumes only 500 µA of power supply

current and features a standby mode which requires only 10 µA,

making it ideal for battery-powered or loop-powered instru-

ments. The part comes in two versions, the AD7714-5, which is

specified for operation from a nominal +5␣ V analog supply

(AV

), and the AD7714-3, which is specified for operation

from a nominal +3.3␣ V analog supply. Both versions can be

operated with a digital supply (DV

) voltage of either +3.3␣ V

or +5␣ V. AD7714Y grade parts operate with a nominal AV

of 3 V or 5 V and can be operated with a digital supply voltage

of either 3 V or 5 V.

The part contains three programmable-gain fully differential

analog input channels that can be reconfigured as five pseudo-

differential inputs. The gain range on all channels is from 1 to

128, allowing the part to accept unipolar signals of between

0 mV to +20␣ mV and 0 V to +2.5␣ V. In bipolar mode, the part

handles genuine bipolar signals of ±20 mV and quasi-bipolar

signals up to ±2.5 V when the reference input voltage equals

+2.5␣ V. With a reference voltage of +1.25␣ V, the input ranges

are from 0 mV to +10 mV to 0 V to +1.25␣ V in unipolar mode,

while in bipolar mode, the part handles genuine bipolar signals

of ±10 mV and quasi-bipolar signals up to ±1.25 V.

The part employs a sigma-delta conversion technique to realize

up to 24 bits of no missing codes performance. The sigma-delta

modulator converts the sampled input signal into a digital pulse

train whose duty cycle contains the digital information. The

programmable gain function on the analog input is also

incorporated in this sigma-delta modulator with the input sam-

pling frequency of the modulator being modified to give the

higher gains. A sinc

digital low-pass filter processes the output

of the sigma-delta modulator and updates the output register at

a rate determined by the first notch frequency of this filter. The

output data can be read from the serial port randomly or peri-

odically at any rate up to the output register update rate. The

first notch of this digital filter, its –3␣ dB frequency and its out-

put rate can be programmed via the filter high and filter low

registers. With a master clock frequency of 2.4576 MHz, the

programmable range for this first notch frequency and output

rate is from 4.8␣ Hz to 1.01 kHz giving a programmable range

for the –3␣ dB frequency of 1.26 Hz to 265␣ Hz.

The basic connection diagram for the part is shown in Figure 2.

This shows both the AV

and DV

pins of the AD7714 being

driven from the analog +3␣ V or +5␣ V supply. Some applications

will have AV

and DV

driven from separate supplies. In the

connection diagram shown, the AD7714’s analog inputs are

configured as three fully differential inputs. The part is set up

for unbuffered mode on the these analog inputs. An AD780,

precision +2.5 V reference, provides the reference source for the

part. On the digital side, the part is configured for three-wire

operation with CS tied to DGND. A quartz crystal or ceramic

resonator provides the master clock source for the part. It may

be necessary to connect capacitors on the crystal or resonator to

ensure that it does not oscillate at overtones of its fundamental

operating frequency. The values of capacitors will vary depend-

ing on the manufacturer’s specifications.

SCLK

MCLK IN

DGND

DVDD

SYNC

RESET

DRDY

MCLK OUT

POL

DIN

DOUT

AIN1

AGND

AIN2

AIN6

AIN3

AIN5

AIN4

REF IN(+)

STANDBY

REF IN(–)

AVDD

BUFFER

AD7714

0.1mF0.1mF

10mF

ANALOG

GROUND

DIFFERENTIAL

ANALOG INPUT 3

DIFFERENTIAL

ANALOG INPUT 2

DIFFERENTIAL

ANALOG INPUT 1

DIGITAL

GROUND

0.1mF10mF

VOUT

VIN

GND

AD780

ANALOG

+5V SUPPLY

ANALOG

+5V SUPPLY

DATA

READY

RECEIVE

(READ)

SERIAL

DATA

SERIAL

CLOCK

CRYSTAL OR

CERAMIC

RESONATOR

+5V

Figure 2. Basic Connection Diagram

AD7714

REV. C–20–

ANALOG INPUT

Analog Input Ranges

The AD7714 contains six analog input pins (labelled AIN1 to

AIN6) which can be configured as either three fully differential

input channels or five pseudo-differential input channels. Bits

CH0, CH1 and CH2 of the Communications Register configure

the analog input arrangement and the channel selection is as

outlined previously in Table VII. The input pairs (either differ-

ential or pseudo-differential) provide programmable-gain, input

channels which can handle either unipolar or bipolar input

signals. It should be noted that the bipolar input signals are

referenced to the respective AIN(–) input of the input pair.

In unbuffered mode, the common-mode range of these inputs is

from AGND to AV

provided that the absolute value of the analog

input voltage lies between AGND␣ –␣ 30␣ mV and AV

+ 30␣ mV.

This means that in unbuffered mode the part can handle both

unipolar and bipolar input ranges for all gains. In buffered

mode, the analog inputs can handle much larger source imped-

ances, but the absolute input voltage range is restricted to be-

tween AGND␣ + 50␣ mV to AV

– 1.5␣ V which also places

restrictions on the common-mode range. This means that in

buffered mode there are some restrictions on the allowable gains

for bipolar input ranges. Care must be taken in setting up the

common-mode voltage and input voltage range so that the

above limits are not exceeded, otherwise there will be a degrada-

tion in linearity performance.

In unbuffered mode, the analog inputs look directly into the

7␣ pF input sampling capacitor, C

SAMP

. The dc input leakage

current in this unbuffered mode is 1␣ nA maximum. As a result,

the analog inputs see a dynamic load which is switched at the

input sample rate (see Figure 3). This sample rate depends on

master clock frequency and selected gain. C

SAMP

is charged to

AIN(+) and discharged to AIN(–) every input sample cycle.

The effective on-resistance of the switch, R

, is typically 7␣ kΩ.

SAMP

must be charged through R

and through any external

source impedances every input sample cycle. Therefore, in unbuf-

fered mode, source impedances mean a longer charge time for

SAMP

and this may result in gain errors on the part. Table XII

shows the allowable external resistance/capacitance values, for

unbuffered mode, such that no gain error to the 16-bit level is

introduced on the part. Table XIII shows the allowable external

resistance/capacitance values, once again for unbuffered mode,

such that no gain error to the 20-bit level is introduced.

Table XII. External R, C Combination for No 16-Bit Gain

Error (Unbuffered Mode Only)

Gain External Capacitance (pF)

0 50 100 500 1000 5000

1 368 kΩ90.6 kΩ54.2 kΩ14.6 kΩ8.2 kΩ2.2 kΩ

2 177.2 kΩ44.2 kΩ26.4 kΩ7.2 kΩ4 kΩ1.12 kΩ

4 82.8 kΩ21.2 kΩ12.6 kΩ3.4 kΩ1.94 kΩ540 Ω

8–128 35.2 kΩ9.6 kΩ5.8 kΩ1.58 kΩ880 Ω240 Ω

Table XIII. External R, C Combination for No 20-Bit Gain

Error (Unbuffered Mode Only)

Gain External Capacitance (pF)

0 50 100 500 1000 5000

1 290 kΩ69 kΩ40.8 kΩ10.4 kΩ5.6 kΩ1.4 kΩ

2 141 kΩ33.8 kΩ20 kΩ5 kΩ2.8 kΩ700 Ω

4 63.6 kΩ16 kΩ9.6 kΩ2.4 kΩ1.34 kΩ340 Ω

8–128 26.8 kΩ7.2 kΩ4.4 kΩ1.1 kΩ600 Ω160 Ω

In buffered mode, the analog inputs look into the high impedance

inputs stage of the on-chip buffer amplifier. C

SAMP

is charged via

this buffer amplifier such that source impedances do not affect

the charging of C

SAMP

. This buffer amplifier has an offset leak-

age current of 1␣ nA. In this buffered mode, large source imped-

ances result in a dc offset voltage developed across the source

impedance but not in a gain error.

Input Sample Rate

The modulator sample frequency for the AD7714 remains at

CLK␣ IN

/128 (19.2␣ kHz @ f

CLK IN

= 2.4576␣ MHz) regardless of

the selected gain. However, gains greater than 1 are achieved

by a combination of multiple input samples per modulator cycle

and a scaling of the ratio of reference capacitor to input capaci-

tor. As a result of the multiple sampling, the input sample rate

of the device varies with the selected gain (see Table XIV). In

buffered mode, the input is buffered before the input sampling

capacitor. In unbuffered mode, where the analog input looks

directly into the sampling capacitor, the effective input imped-

ance is 1/C

SAMP

× f

where C

SAMP

is the input sampling capaci-

tance and f

is the input sample rate.

RSW (7kV TYP) HIGH

IMPEDANCE

>1GV

CSAMP

(7pF )

VBIAS

SWITCHING FREQUENCY DEPENDS ON

fCLKIN AND SELECTED GAIN

AIN(+)

AIN(–)

Figure 3. Unbuffered Analog Input Structure

AD7714

REV. C –21–

Table XIV. Input Sampling Frequency vs. Gain

Gain Input Sampling Freq (f

)

CLK IN

/64 (38.4␣ kHz @ f

CLK IN

= 2.4576␣ MHz)

22 × f

CLK IN

/64 (76.8␣ kHz @ f

CLK IN

= 2.4576␣ MHz)

44 × f

CLK IN

/64 (153.6␣ kHz @ f

CLK IN

= 2.4576␣ MHz)

88 × f

CLK IN

/64 (307.2␣ kHz @ f

CLK IN

= 2.4576␣ MHz)

16 8 × f

CLK IN

/64 (307.2␣ kHz @ f

CLK IN

= 2.4576␣ MHz)

32 8 × f

CLK IN

/64 (307.2␣ kHz @ f

CLK IN

= 2.4576␣ MHz)

64 8 × f

CLK IN

/64 (307.2␣ kHz @ f

CLK IN

= 2.4576␣ MHz)

128 8 × f

CLK IN

/64 (307.2␣ kHz @ f

CLK IN

= 2.4576␣ MHz)

Burnout Current

The AD7714 contains two 1␣ µA currents, one source current

from AV

to AIN(+) and one sink from AIN(–) to AGND. The

currents are either both on or off depending on the BO bit of the

Mode Register. These currents can be used in checking that a

transducer has not burned out nor gone open-circuit before

attempting to take measurements on that channel. If the cur-

rents are turned on, allowed flow in the transducer, a measure-

ment of the input voltage on the analog input taken and the

voltage measured is full scale, it indicates that the transducer has

gone open-circuit; if the voltage measured is zero, it indicates

that the transducer has gone short-circuit. For normal opera-

tion, these burnout currents are turned off by writing a 0 to the

BO bit. For the source current to work correctly, the applied

voltage on AIN(+) should not go within 500␣ mV of AV

. For

the sink current to work correctly, the applied voltage on the

AIN(–) input should not go within 500␣ mV of AGND.

Bipolar/Unipolar Inputs

The analog inputs on the AD7714 can accept either unipolar or

bipolar input voltage ranges. Bipolar input ranges do not imply

that the part can handle negative voltages on its analog inputs,

since the analog input cannot go more negative than –30␣ mV to

ensure correct operation of the part. The input channels are

either fully differential or pseudo-differential (all other channels

referenced to AIN6). In either case, the input channels are

arranged in pairs with an AIN(+) and AIN(–). As a result, the

voltage to which the unipolar and bipolar signals on the AIN(+)

input are referenced is the voltage on the respective AIN(–)

input. For example, if AIN(–) is +2.5␣ V and the AD7714 is

configured for unipolar operation with a gain of 2 and a V

REF

+2.5␣ V, the input voltage range on the AIN(+) input is +2.5 V to

+3.75␣ V. If AIN(–) is +2.5␣ V and the AD7714 is configured for

bipolar mode with a gain of 2 and a V

REF

of +2.5␣ V, the analog

input range on the AIN(+) input is +1.25␣ V to +3.75 V (i.e.,

2.5␣ V ± 1.25␣ V). If AIN(–) is at AGND, the part cannot be con-

figured for bipolar ranges in excess of ±30␣ mV.

Bipolar or unipolar options are chosen by programming the B/U

bit of the Filter High Register. This programs the selected chan-

nel for either unipolar or bipolar operation. Programming the

channel for either unipolar or bipolar operation does not change

any of the input signal conditioning; it simply changes the data

output coding and the points on the transfer function where

calibrations occur.

REFERENCE INPUT

The AD7714’s reference inputs, REF␣ IN(+) and REF␣ IN(–),

provide a differential reference input capability. The common-

mode range for these differential inputs is from AGND to AV

The nominal reference voltage, V

REF

(REF␣ IN(+)␣ –REF␣ IN(–)),

for specified operation is +2.5␣ V for the AD7714-5 and +1.25␣ V

for the AD7714-3. The part is functional with V

REF

voltages

down to 1 V but with degraded performance as the output noise

will, in terms of LSB size, be larger. REF␣ IN(+) must always be

greater than REF␣ IN(–) for correct operation of the AD7714.

Both reference inputs provide a high impedance, dynamic load

similar to the analog inputs in unbuffered mode. The maxi-

mum dc input leakage current is ±1 nA over temperature and

source resistance may result in gain errors on the part. In this

case, the sampling switch resistance is 5␣ kΩ typ and the refer-

ence capacitor (C

REF

) varies with gain. The sample rate on the

reference inputs is f

CLK IN

/64 and does not vary with gain. For

gains of 1 to 8, C

REF

is 8 pF; for a gain of 16, it is 5.5 pF, for a

gain of 32, it is 4.25 pF, for a gain of 64, it is 3.625 pF and for a

gain of 128, it is 3.3125 pF.

The output noise performance outlined in Tables I through IV

is for an analog input of 0 V and is unaffected by noise on the

reference. To obtain the same noise performance as shown in

the noise tables over the full input range requires a low noise

reference source for the AD7714. If the reference noise in the

bandwidth of interest is excessive, it will degrade the perfor-

mance of the AD7714. In applications where the excitation

voltage for the bridge transducer on the analog input also de-

rives the reference voltage for the part, the effect of the noise in

the excitation voltage will be removed as the application is

ratiometric. Recommended reference voltage sources for the

AD7714-5 and AD7714Y grade with AV

= 5 V include the

AD780, REF43 and REF192 while the recommended reference

sources for the AD7714-3 and AD7714Y with AV

= 3 V

include the AD589 and AD1580. It is generally recommended

to decouple the output of these references to further reduce the

noise level.

DIGITAL FILTERING

The AD7714 contains an on-chip low-pass digital filter which

processes the output of the part’s sigma-delta modulator. There-

fore, the part not only provides the analog-to-digital conversion

function but it also provides a level of filtering. There are a

number of system differences when the filtering function is

provided in the digital domain rather than the analog domain

and the user should be aware of these.

First, since digital filtering occurs after the A-to-D conversion

process, it can remove noise injected during the conversion

process. Analog filtering cannot do this. Also, the digital filter

can be made programmable far more readily than an analog

filter. Depending on the digital filter design, this gives the user

the capability of programming cutoff frequency and output

update rate.

On the other hand, analog filtering can remove noise superim-

posed on the analog signal before it reaches the ADC. Digital

filtering cannot do this and noise peaks riding on signals near

full scale have the potential to saturate the analog modulator

and digital filter, even though the average value of the signal is

within limits. To alleviate this problem, the AD7714 has over-

range headroom built into the sigma-delta modulator and digital

filter which allows overrange excursions of 5% above the analog

input range. If noise signals are larger than this, consideration

should be given to analog input filtering, or to reducing the

input channel voltage so that its full scale is half that of the

analog input channel full scale. This will provide an overrange

capability greater than 100% at the expense of reducing the

dynamic range by 1 bit (50%).

AD7714

REV. C–22–

The cutoff frequency of the digital filter is determined by the

value loaded to bits FS0 to FS11 in the Filter High and Filter

Low Registers. Programming a different cutoff frequency via

FS0 – FS11 does not alter the profile of the filter response; it

changes the frequency of the notches as outlined in the Filter

Registers section. The output update and first notch correspond

and are determined by the relationship:

Output Rate = f

CLK IN

/(N.128)

where N is the decimal equivalent of the word loaded to the

FS0 to FS11 bits of the Filter Registers

while the –3␣ dB frequency is determined by the relationship:

–3␣ dB frequency = 0.262

filter first notch frequency

The filter provides a linear phase response with a group delay

determined by:

Group Delay = –3

.(N.f/f

MOD

)

where N is the decimal equivalent of the word loaded to the

FS0 to FS11 bits of the Filter Registers and f

MOD

= f

CLK IN

/128.

Since the AD7714 contains this on-chip, low-pass filtering, a

settling time is associated with step function inputs and data on

the output will be invalid after a step change until the settling

time has elapsed. The settling time depends upon the output

rate chosen for the filter. The settling time of the filter to a full-

scale step input can be up to four times the output data period.

For a synchronized step input (using the SYNC or FSYNC

functions) the settling time is three times the output data pe-

riod. When changing channels on the part, the change from one

channel to the other is synchronized so the output settling time

is also three times the output data period. Thus, in switching

between channels, the output data register is not updated until

the settling time of the filter has elapsed.

Post-Filtering

The on-chip modulator provides samples at a 19.2␣ kHz output

rate with f

CLK IN

at 2.4576␣ MHz. The on-chip digital filter

decimates these samples to provide data at an output rate that

corresponds to the programmed output rate of the filter. Since

the output data rate is higher than the Nyquist criterion, the

output rate for a given bandwidth will satisfy most application

requirements. However, there may be some applications that

require a higher data rate for a given bandwidth and noise per-

formance. Applications that need this higher data rate will

require some post-filtering following the part’s digital filter.

For example, if the required bandwidth is 7.86␣ Hz but the

required update rate is 100␣ Hz, the data can be taken from the

AD7714 at the 100␣ Hz rate giving a –3 dB bandwidth of

26.2␣ Hz. Post-filtering can be applied to this to reduce the

bandwidth and output noise, to the 7.86␣ Hz bandwidth level,

while maintaining an output rate of 100␣ Hz.

Post-filtering can also be used to reduce the output noise from

the device for bandwidths below 1.26␣ Hz. At a gain of 128 and

a bandwidth of 1.26␣ Hz, the output rms noise is 140␣ nV. This

is essentially device noise or white noise and since the input is

chopped, the noise has a primarily flat frequency response. By

reducing the bandwidth below 1.26␣ Hz, the noise in the result-

ant passband can be reduced. A reduction in bandwidth by a

factor of 2 results in a reduction of approximately 1.25 in the

output rms noise. This additional filtering will result in a

longer settling time.

In addition, the digital filter does not provide any rejection at

integer multiples of the digital filter’s sample frequency. How-

ever, the input sampling on the part provides attenuation at

multiples of the digital filter’s sampling frequency so that the

unattenuated bands actually occur around multiples of the input

sampling frequency f

(as defined in Table XIV). Thus, the

unattenuated bands occur at n × f

(where n = 1, 2, 3. . .). At

these frequencies, there are frequency bands, ±f

3 dB

wide (f

3 dB

the cutoff frequency of the digital filter) at either side where

noise passes unattenuated to the output.

Filter Characteristics

The AD7714’s digital filter is a low-pass filter with a (sinx/x)

response (also called sinc

). The transfer function for this filter

is described in the z-domain by:

H(z)=1

N×1−Z−N

1−Z−1













and in the frequency domain by:

Hf N

Sin N f f

Sin f f

() (.. )

(. )

=×

Figure 4 shows the filter frequency response for a cutoff

frequency of 2.62␣ Hz which corresponds to a first filter notch

frequency of 10␣ Hz. The plot is shown from dc to 65␣ Hz.

This response is repeated at either side of the input sampling

frequency and at either side of multiples of the input sampling

frequency.

FREQUENCY – Hz

600

–40

50302010 40

–60

–80

–100

–120

–140

–160

–180

–200

–220

–20

–240

GAIN – dB

Figure 4. Frequency Response of AD7714 Filter

The response of the filter is similar to that of an averaging filter

but with a sharper roll-off. The output rate for the digital filter

corresponds with the positioning of the first notch of the filter’s

frequency response. Thus, for the plot of Figure 4 where the

output rate is 10␣ Hz, the first notch of the filter is at 10␣ Hz. The

notches of this (sinx/x)

filter are repeated at multiples of the

first notch. The filter provides attenuation of better than 100 dB

at these notches. For the example given, if the first notch is at

10␣ Hz, there will be notches (and hence >100␣ dB rejection) at

both 50␣ Hz and 60␣ Hz.

AD7714

REV. C –23–

value which, when normalized, is subtracted from all conversion

results. The full-scale calibration register contains a value

which, when normalized, is multiplied by all conversion results.

The offset calibration coefficient is subtracted from the result

prior to the multiplication by the full-scale coefficient. This

means that the full-scale coefficient is effectively a span or gain

coefficient.

The AD7714 offers self-calibration, system calibration and

background calibration facilities. For full calibration to occur

on the selected channel, the on-chip microcontroller must record

the modulator output for two different input conditions. These

are “zero-scale” and “full-scale” points. These points are de-

rived by performing a conversion on the different input voltages

provided to the input of the modulator during calibration. As a

result, the accuracy of the calibration can only be as good as the

noise level which the part provides in normal mode. The result

of the “zero-scale” calibration conversion is stored in the Zero

Scale Calibration Register for the appropriate channel. The

result of the “full-scale” calibration conversion is stored in the

Full-Scale Calibration Register for the appropriate channel. With

these readings, the microcontroller can calculate the offset and

the gain slope for the input to output transfer function of the

converter. Internally, the part works with 33 bits of resolution

to determine its conversion result of either 16 bits or 24 bits.

Self-Calibration

A self-calibration is initiated on the AD7714 by writing the

appropriate values (0, 0, 1) to the MD2, MD1 and MD0 bits of

the Mode Register. In the self-calibration mode with a unipolar

input range, the zero-scale point used in determining the cali-

bration coefficients is with the inputs of the differential pair

internally shorted on the part (i.e., AIN(+) = AIN(–) = Internal

Bias Voltage). The PGA is set for the selected gain (as per G2,

G1, G0 bits in the Mode Register) for this zero-scale calibration

conversion. The full-scale calibration conversion is performed at

the selected gain on an internally-generated voltage of V

REF

Selected Gain.

The duration time of the calibration is 6 × 1/Output Rate. This

is made up of 3 × 1/Output Rate for the zero-scale calibration

and 3 × 1/Output Rate for the full-scale calibration. At this time

the MD2, MD1 and MD0 bits in the Mode Register return to

0, 0, 0. This gives the earliest indication that the calibration

sequence is complete. The DRDY line goes high when calibra-

tion is initiated and does not return low until there is a valid

new word in the data register. The duration time from the cali-

bration command being issued to DRDY going low is 9 × 1/

Output Rate. This is made up of 3 × 1/Output Rate for the zero-

scale calibration, 3 × 1/Output Rate for the full-scale calibration

and 3 × 1/Output Rate for a conversion on the analog input. If

DRDY is low before (or goes low during) the calibration com-

mand write to the Mode Register, it may take up to one modu-

lator cycle (MCLK␣ IN/128) before DRDY goes high to indicate

that calibration is in progress. Therefore, DRDY should be

ignored for up to one modulator cycle after the last bit of the

calibration command is written to the Mode Register.

For bipolar input ranges in the self-calibrating mode, the se-

quence is very similar to that just outlined. In this case, the two

points are exactly the same as above but since the part is config-

ured for bipolar operation, the output code for zero differential

input is 800000 Hex in 24-bit mode.

ANALOG FILTERING

The digital filter does not provide any rejection at integer mul-

tiples of the input sampling frequency, as outlined earlier. How-

ever, due to the AD7714’s high oversampling ratio, these bands

occupy only a small fraction of the spectrum and most broad-

band noise is filtered. This means that the analog filtering re-

quirements in front of the AD7714 are considerably reduced

versus a conventional converter with no on-chip filtering. In

addition, because the part’s common-mode rejection perfor-

mance of 100␣ dB extends out to several kHz, common-mode

noise in this frequency range will be substantially reduced.

Depending on the application, however, it may be necessary to

provide attenuation prior to the AD7714 in order to eliminate

unwanted frequencies from these bands which the digital filter

will pass. It may also be necessary in some applications to pro-

vide analog filtering in front of the AD7714 to ensure that dif-

ferential noise signals outside the band of interest do not

saturate the analog modulator.

If passive components are placed in front of the AD7714, in

unbuffered mode, care must be taken to ensure that the source

impedance is low enough so as not to introduce gain errors in

the system. This significantly limits the amount of passive anti-

aliasing filtering which can be provided in front of the AD7714

when it is used in unbuffered mode. However, when the part is

used in buffered mode, large source impedances will simply

result in a small dc offset error (a 10␣ kΩ source resistance will

cause an offset error of less than 10␣ µV). Therefore, if the sys-

tem requires any significant source impedances to provide pas-

sive analog filtering in front of the AD7714, it is recommended

that the part be operated in buffered mode.

CALIBRATION

The AD7714 provides a number of calibration options which

can be programmed via the MD2, MD1 and MD0 bits of the

Mode Register. The different calibration options are outlined

in the Mode Register and Calibration Sequences sections. A

calibration cycle may be initiated at any time by writing to these

bits of the Mode Register. Calibration on the AD7714 removes

offset and gain errors from the device. A calibration routine

should be initiated on the device whenever there is a change in

the ambient operating temperature or supply voltage. It should

also be initiated if there is a change in the selected gain, filter

notch or bipolar/unipolar input range.

The AD7714 gives the user access to the on-chip calibration

registers allowing the microprocessor to read the device’s cali-

bration coefficients and also to write its own calibration coeffi-

cients to the part from prestored values in E

PROM. This gives

the microprocessor much greater control over the AD7714’s

calibration procedure. It also means that the user can verify

that the device has performed its calibration correctly by com-

paring the coefficients after calibration with prestored values in

PROM. The values in these calibration registers are 24-bit

wide. In addition, the span and offset for the part can be

adjusted by the user.

There is a significant variation in the value of these coefficients

across the different output update rates, gains and unipolar/

bipolar operation. Internally in the AD7714, these coefficients

are normalized before being used to scale the words coming out

of the digital filter. The offset calibration register contains a

AD7714

REV. C–24–

The part also offers ZS Self-Calibration and FS Self-Calibration

options. In these cases, the part performs just a zero-scale or

full-scale calibration respectively and not a full calibration of the

part. A full-scale calibration should not be carried out unless

the part contains valid zero-scale coefficients. These calibrations

are initiated on the AD7714 by writing the appropriate values

(1, 1, 0 for ZS Self-Calibration and 1, 1, 1 for FS Self Calibra-

tion) to the MD2, MD1 and MD0 bits of the Mode Register.

The zero-scale or full-scale calibration is exactly the same as

that described for the full self-calibration. In these cases, the

duration of the calibration is 3 × 1/Output Rate. At this time the

MD2, MD1 and MD0 bits in the Mode Register return to

0, 0, 0. This gives the earliest indication that the calibration

sequence is complete. The DRDY line goes high when calibra-

tion is initiated and does not return low until there is a valid

new word in the data register. The time from the calibration

command being issued to DRDY going low is 6 × 1/Output

Rate. This is made up of 3 × 1/Output Rate for the zero-scale or

full-scale calibration and 3 × 1/Output Rate for a conversion on

the analog input. If DRDY is low before (or goes low during)

the calibration command write to the Mode Register, it may

take up to one modulator cycle (MCLK␣ IN/128) before DRDY

goes high to indicate that calibration is in progress. Therefore,

DRDY should be ignored for up to one modulator cycle after

the last bit of the calibration command is written to the Mode

The fact that the self-calibration can be performed as a two step

calibration offers another feature. After the sequence of a full

self calibration has been completed, additional offset or gain

calibrations can be performed by themselves to adjust the part’s

zero point or gain. Calibrating one of the parameters, either

offset or gain, will not affect the other parameter.

System Calibration

System calibration allows the AD7714 to compensate for system

gain and offset errors as well as its own internal errors. System

calibration performs the same slope factor calculations as self-

calibration but uses voltage values presented by the system to

the AIN inputs for the zero- and full-scale points. Full System

calibration requires a two-step process, a ZS System Calibration

followed by a FS System Calibration.

For a full system calibration, the zero-scale point must be pre-

sented to the converter first. It must be applied to the converter

before the calibration step is initiated and remain stable until the

step is complete. Once the system zero scale has been set up at

the analog input, a ZS System Calibration is then initiated by

writing the appropriate values (0, 1, 0) to the MD2, MD1 and

MD0 bits of the Mode Register. The zero-scale system calibra-

tion is performed at the selected gain. The duration of the cali-

bration is 3 × 1/Output Rate. At this time, the MD2, MD1 and

MD0 bits in the Mode Register return to 0, 0, 0. This gives the

earliest indication that the calibration sequence is complete. The

DRDY line goes high when calibration is initiated and does not

return low until there is a valid new word in the data register.

The time from the calibration command being issued to DRDY

going low is 4 × 1/Output Rate. This is made up of 3 × 1/Output

Rate for the zero-scale system calibration and 1/Output Rate for

a conversion on the analog input. This conversion on the analog

input is on the same voltage as the zero-scale system calibration

and, therefore, the resultant word in the data register from this

conversion should be a zero-scale reading. If DRDY is low

before (or goes low during) the calibration command write to

the Mode Register, it may take up to one modulator cycle

(MCLK␣ IN/128) before DRDY goes high to indicate that cali-

bration is in progress. Therefore, DRDY should be ignored for

up to one modulator cycle after the last bit of the calibration

command is written to the Mode Register.

After the zero-scale point is calibrated, the full-scale point is

applied to AIN and the second step of the calibration process is

initiated by again writing the appropriate values (0, 1, 1) to

MD2, MD1 and MD0. Again the full-scale voltage must be set

up before the calibration is initiated, and it must remain stable

throughout the calibration step. The full-scale system calibra-

tion is performed at the selected gain. The duration of the cali-

bration is 3 × 1/Output Rate. At this time, the MD2, MD1 and

MD0 bits in the Mode Register return to 0, 0, 0. This gives the

earliest indication that the calibration sequence is complete. The

DRDY line goes high when calibration is initiated and does not

return low until there is a valid new word in the data register.

The time from the calibration command being issued to DRDY

going low is 4 × 1/Output Rate. This is made up of 3 × 1/Out-

put Rate for the full-scale system calibration and 1/Output Rate

for a conversion on the analog input. This conversion on the

analog input is on the same voltage as the full-scale system

calibration and, therefore, the resultant word in the data register

from this conversion should be a full-scale reading. If DRDY is

low before (or goes low during) the calibration command write

to the Mode Register, it may take up to one modulator cycle

(MCLK␣ IN/128) before DRDY goes high to indicate that cali-

bration is in progress. Therefore, DRDY should be ignored for

up to one modulator cycle after the last bit of the calibration

command is written to the Mode Register.

In the unipolar mode, the system calibration is performed

between the two endpoints of the transfer function; in the bipo-

lar mode, it is performed between midscale (zero differential

voltage) and positive full scale.

The fact that the system calibration is a two step calibration

offers another feature. After the sequence of a full system cali-

bration has been completed, additional offset or gain calibra-

tions can be performed by themselves to adjust the system zero

reference point or the system gain. Calibrating one of the

parameters, either system offset or system gain, will not affect

the other parameter. A full-scale calibration should not be car-

ried out unless the part contains valid zero-scale coefficients.

System calibration can also be used to remove any errors from

source impedances on the analog input when the part is used in

unbuffered mode. A simple R, C antialiasing filter on the front

end may introduce a gain error on the analog input voltage but

the system calibration can be used to remove this error.

AD7714

REV. C –25–

System-Offset Calibration

System-offset calibration is a variation of both the system cali-

bration and self-calibration. In this case, the zero-scale point is

determined in exactly the same way as a ZS System Calibration.

The system zero-scale point is presented to the AIN inputs of

the converter. This must be applied to the converter before

the calibration step is initiated and remain stable until the step

is complete. Once the system zero scale has been set up, a

System-Offset Calibration is then initiated by writing the appro-

priate values (1, 0, 0) to the MD2, MD1 and MD0 bits of the

Mode Register. The zero-scale system calibration is performed

at the selected gain.

The full-scale calibration is performed in exactly the same way

as an FS Self Calibration. The full-scale calibration conversion

is performed at the selected gain on an internally generated

voltage of V

REF

/Selected Gain. This is a one step calibration

sequence and the time for calibration is 6 ×1/Output Rate. At

this time, the MD2, MD1 and MD0 bits in the Mode Register

return to 0, 0, 0. This gives the earliest indication that the cali-

bration sequence is complete. The DRDY line goes high when

calibration is initiated and does not return low until there is a

valid new word in the data register. The duration time from the

calibration command being issued to DRDY going low is 9 ×1/

Output Rate. This is made up of 3 ×1/Output Rate for the zero-

scale system calibration, 3 ×1/Output Rate for the full-scale

self-calibration and 3 ×1/Output Rate for a conversion on the

analog input. This conversion on the analog input is on the

same voltage as the zero-scale system calibration and, therefore,

the resultant word in the data register from this conversion

should be a zero-scale reading. If DRDY is low before (or goes

low during) the calibration command write to the Mode Regis-

ter, it may take up to one modulator cycle (MCLK␣ IN/128)

before DRDY goes high to indicate that calibration is in

progress. Therefore, DRDY should be ignored for up to one

modulator cycle after the last bit of the calibration command is

written to the Mode Register.

In the unipolar mode, the system-offset calibration is performed

between the two endpoints of the transfer function; in the bipolar

mode, it is performed between midscale and positive full scale.

Background Calibration

The AD7714 also offers a background calibration mode where

the part interleaves its calibration procedure with its normal

conversion sequence. In the background calibration mode, the

part provides continuous zero-scale self-calibrations; it does not

provide any full-scale calibrations. The zero-scale point used in

determining the calibration coefficients in this mode is exactly

the same as for a ZS Self-Calibration. The background calibra-

tion mode is invoked by writing 1, 0, 1 to the MD2, MD1,

MD0 bits of the Mode Register. When invoked, the back-

ground calibration mode performs a zero-scale self calibration

after every output update and this reduces the output data rate

of the AD7714 by a factor of six. Its advantage is that the part

is continually performing offset calibrations and automatically

updating its zero-scale calibration coefficients. As a result, the

effects of temperature drift, supply sensitivity and time drift on

zero-scale errors are automatically removed. When the back-

ground calibration mode is turned on, the part will remain in

this mode until bits MD2, MD1 and MD0 of the Mode Regis-

ter are changed.

Because the background calibration does not perform full-scale

calibrations, a self-calibration should be performed before plac-

ing the part in background calibration mode. Removal of the

offset drift in this mode leaves gain drift as the only source of

error not removed from the part. The typical gain drift of the

AD7714 with temperature is 0.2␣ ppm/°C. The SYNC input or

FSYNC bit should not be exercised when the part is in back-

ground calibration mode.

Span and Offset Limits

Whenever a system calibration mode is used, there are limits on

the amount of offset and span which can be accommodated.

The overriding requirement in determining the amount of offset

and gain which can be accommodated by the part is the require-

ment that the positive full-scale calibration limit is ≤ 1.05 ×

REF

/GAIN. This allows the input range to go 5% above the

nominal range. The built-in headroom in the AD7714’s analog

modulator ensures that the part will still operate correctly with a

positive full-scale voltage which is 5% beyond the nominal.

The range of input span in both the unipolar and bipolar modes

has a minimum value of 0.8 ×V

REF

/GAIN and a maximum

value of 2.1 ×V

REF

/GAIN. However, the span (which is the

difference between the bottom of the AD7714’s input range and

the top of its input range) has to take into account the limitation

on the positive full-scale voltage. The amount of offset which

can be accommodated depends on whether the unipolar or

bipolar mode is being used. Once again, the offset has to take

into account the limitation on the positive full-scale voltage. In

unipolar mode, there is considerable flexibility in handling nega-

tive (with respect to AIN(–)) offsets. In both unipolar and bipo-

lar modes, the range of positive offsets which can be handled by

the part depends on the selected span. Therefore, in determin-

ing the limits for system zero-scale and full-scale calibrations,

the user has to ensure that the offset range plus the span range

does exceed 1.05 ×V

REF

/GAIN. This is best illustrated by

looking at a few examples.

If the part is used in unipolar mode with a required span of

0.8 ×V

REF

/GAIN, the offset range the system calibration can

handle is from –1.05 ×V

REF

/GAIN to +0.25 ×V

REF

/GAIN. If

the part is used in unipolar mode with a required span of V

REF

GAIN, the offset range the system calibration can handle is

from –1.05 ×V

REF

/GAIN to +0.05 ×V

REF

/GAIN. Similarly, if

the part is used in unipolar mode and required to remove an

offset of 0.2 ×V

REF

/GAIN, the span range the system calibra-

tion can handle is 0.85 ×V

REF

/GAIN.

If the part is used in bipolar mode with a required span of

±0.4 ×V

REF

/GAIN, then the offset range which the system cali-

bration can handle is from –0.65 ×V

REF

/GAIN to +0.65 ×

REF

/GAIN. If the part is used in bipolar mode with a required

span of ±V

REF

/GAIN, the offset range the system calibration can

handle is from –0.05 ×V

REF

/GAIN to +0.05 × V

REF

/GAIN.

Similarly, if the part is used in bipolar mode and required to

remove an offset of ±0.2 ×V

REF

/GAIN, the span range the sys-

tem calibration can handle is ±0.85 × V

REF

/GAIN.

AD7714

REV. C–26–

Power-Up and Calibration

On power-up, the AD7714 performs an internal reset which sets

the contents of the internal registers to a known state. There

are default values loaded to all registers after a power-on or

reset. The default values contain nominal calibration coefficients

for the calibration registers. However, to ensure correct calibra-

tion for the device a calibration routine should be performed

after power-up.

The power dissipation and temperature drift of the AD7714 are

low and no warm-up time is required before the initial calibra-

tion is performed. However, if an external reference is being

used, this reference must have stabilized before calibration is

initiated. Similarly, if the clock source for the part is generated

from a crystal or resonator across the MCLK pins, the start-up

time for the oscillator circuit should elapse before a calibration

is initiated on the part (see below).

USING THE AD7714

Clocking and Oscillator Circuit

The AD7714 requires a master clock input, which may be an

external CMOS compatible clock signal applied to the MCLK␣ IN

pin with the MCLK␣ OUT pin left unconnected. Alternatively, a

crystal or ceramic resonator of the correct frequency can be

connected between MCLK␣ IN and MCLK␣ OUT in which case

the clock circuit will function as an oscillator, providing the

clock source for the part. The input sampling frequency, the

modulator sampling frequency, the –3␣ dB frequency, output

update rate and calibration time are all directly related to the

master clock frequency, f

CLK␣ IN

. Reducing the master clock

frequency by a factor of 2 will halve the above frequencies and

update rate and double the calibration time. The current drawn

from the DV

power supply is also directly related to f

CLK␣ IN

Reducing f

CLK␣ IN

by a factor of 2 will halve the DV

current

but will not affect the current drawn from the AV

power supply.

Using the part with a crystal or ceramic resonator between the

MCLK IN and MCLK OUT pins generally causes more cur-

rent to be drawn from DV

than when the part is clocked from

a driven clock signal at the MCLK IN pin. This is because the

on-chip oscillator circuit is active in the case of the crystal or

ceramic resonator. Therefore, the lowest possible current on

the AD7714 is achieved with an externally applied clock at the

MCLK IN pin with MCLK OUT unconnected and unloaded.

The amount of additional current taken by the oscillator

depends on a number of factors—first, the larger the value of

capacitor placed on the MCLK␣ IN and MCLK␣ OUT pins, then

the larger the DV

current consumption on the AD7714. Care

should be taken not to exceed the capacitor values recommended

by the crystal and ceramic resonator manufacturers to avoid

consuming unnecessary DV

current. Typical values recom-

mended by crystal or ceramic resonator manufacturers are in the

range of 30␣ pF to 50␣ pF and if the capacitor values on MCLK

IN and MCLK OUT are kept in this range they will not result

in any excessive DV

current. Another factor that influences

the DV

current is the effective series resistance (ESR) of the

crystal which appears between the MCLK IN and MCLK OUT

pins of the AD7714. As a general rule, the lower the ESR value

then the lower the current taken by the oscillator circuit.

When operating with a clock frequency of 2.4576␣ MHz, there is

no appreciable difference in the DV

current between an

externally applied clock and a crystal resonator when operating

with a DV

of +3␣ V. With DV

= +5␣ V and f

CLK IN

2.4576␣ MHz, the typical DV

current increases by 50␣ µA for a

crystal/resonator supplied clock versus an externally applied

clock. The ESR values for crystals and resonators at this fre-

quency tend to be low and as a result there tends to be little

difference between different crystal and resonator types.

When operating with a clock frequency of 1␣ MHz, the ESR

value for different crystal types varies significantly. As a result,

the DV

current drain varies across crystal types. When using

a crystal with an ESR of 700␣ Ω or when using a ceramic resona-

tor, the increase in the typical DV

current over an externally-

applied clock is 50␣ µA with DV

= +3␣ V and 175␣ µA with

= +5␣ V. When using a crystal with an ESR of 3␣ kΩ, the

increase in the typical DV

current over an externally applied

clock is again 50␣ µA with DV

= +3␣ V but 300␣ µA with

= +5␣ V.

The on-chip oscillator circuit also has a start-up time associated

with it before it is oscillating at its correct frequency and correct

voltage levels. The typical start up time for the circuit is 10␣ ms

with a DV

of +5␣ V and 15␣ ms with a DV

of +3␣ V. At 3␣ V

supplies, depending on the loading capacitances on the MCLK

pins, a 1␣ MΩ feedback resistor may be required across the crys-

tal or resonator in order to keep the start up times around the

15␣ ms duration.

The AD7714’s master clock appears on the MCLK OUT pin of

the device. The maximum recommended load on this pin is one

CMOS load. When using a crystal or ceramic resonator to gen-

erate the AD7714’s clock, it may be desirable to then use this

clock as the clock source for the system. In this case, it is recom-

mended that the MCLK OUT signal is buffered with a CMOS

buffer before being applied to the rest of the circuit.

System Synchronization

The SYNC input (or FSYNC bit) allows the user to reset the

modulator and digital filter without affecting any of the setup

conditions on the part. This allows the user to start gathering

samples of the analog input from a known point in time, i.e., the

rising edge of SYNC or when a 1 is written to FSYNC.

The SYNC input can also be used to allow two other functions.

If multiple AD7714s are operated from a common master clock,

they can be synchronized to update their output registers simul-

taneously. A falling edge on the SYNC input (or a 1 written to

the FSYNC bit of the Mode Register) resets the digital filter and

analog modulator and places the AD7714 into a consistent,

known state. While the SYNC input is low (or FSYNC high),

the AD7714 will be maintained in this state. On the rising edge

of SYNC (or when a 0 is written to the FSYNC bit), the modu-

lator and filter are taken out of this reset state and on the next

clock edge the part starts to gather input samples again. In a

system using multiple AD7714s, a common signal to their

SYNC inputs will synchronize their operation. This would nor-

mally be done after each AD7714 has performed its own cali-

bration or has had calibration coefficients loaded to it. The

output updates will then be synchronized with the maximum

possible difference between the output updates of the individual

AD7714s being one MCLK IN cycle.

AD7714

REV. C –27–

The SYNC input can also be used as a start convert command

allowing the AD7714 to be operated in a conventional converter

fashion. In this mode, the rising edge of SYNC starts conversion

and the falling edge of DRDY indicates when conversion is

complete. The disadvantage of this scheme is that the settling

time of the filter has to be taken into account for every data

ter is updated at a three times slower rate in this mode.

Since the SYNC input (or FSYNC bit) resets the digital filter,

the full settling-time of 3 × 1/Output Rate has to elapse before

there is a new word loaded to the output register on the part. If

the DRDY signal is low when SYNC returns high (or FSYNC

goes to a 0), the DRDY signal will not be reset high by the

SYNC (or FSYNC) command. This is because the AD7714

recognizes that there is a word in the data register which has not

been read. The DRDY line will stay low until an update of the

data register takes place at which time it will go high for

500 ×t

CLK IN

before returning low again. A read from the data

until the settling time of the filter has elapsed (from the SYNC

or FSYNC command) and there is a valid new word in the data

command is issued, the DRDY line will not return low until the

settling time of the filter has elapsed.

Reset Input

The RESET input on the AD7714 resets all the logic, the digital

filter and the analog modulator while all on-chip registers are

reset to their default state. DRDY is driven high and the

AD7714 ignores all communications to any of its registers while

the RESET input is low. When the RESET input returns high,

the AD7714 starts to process data and DRDY will return low in

3×1/Output Rate indicating a valid new word in the data regis-

ter. However, the AD7714 operates with its default setup condi-

tions after a RESET and it is generally necessary to set up all

registers and carry out a calibration after a RESET command.

The AD7714’s on-chip oscillator circuit continues to function

even when the RESET input is low. The master clock signal

continues to be available on the MCLK OUT pin. Therefore, in

applications where the system clock is provided by the AD7714’s

clock, the AD7714 produces an uninterrupted master clock

during RESET commands.

Standby Mode

The STANDBY input on the AD7714 allows the user to place

the part in a power-down mode when it is not required to pro-

vide conversion results The AD7714 retains the contents of all

its on-chip registers (including the data register) while in

standby mode. When in standby mode, the digital interface is

reset and DRDY is reset to a Logic 1. Data cannot be accessed

from the part while in standby mode. When released from standby

mode, the part starts to process data and a new word is available

in the data register in 3 ×1/Output rate from when the STANDBY

input goes high.

Placing the part in standby mode reduces the total current to

5␣ µA typical when the part is operated from an external master

clock, provided this master clock is stopped. If the external

clock continues to run in standby mode, the standby current

increases to 150␣ µA typical with 5 V supplies and 75 µA typical

with 3.3 V supplies. If a crystal or ceramic resonator is used as

the clock source, the total current in standby mode is 400␣ µA

typical with 5 V supplies and 90 µA with 3.3 V supplies. This is

because the on-chip oscillator circuit continues to run when the

part is in its standby mode. This is important in applications

where the system clock is provided by the AD7714’s clock, so

that the AD7714 produces an uninterrupted master clock even

when it is in its standby mode.

Accuracy

Sigma-Delta ADCs, like VFCs and other integrating ADCs, do

not contain any source of nonmonotonicity and inherently offer

no missing codes performance. The AD7714 achieves excellent

linearity by the use of high quality, on-chip capacitors, which

have a very low capacitance/voltage coefficient. The device also

achieves low input drift through the use of chopper-stabilized

techniques in its input stage. To ensure excellent performance

over time and temperature, the AD7714 uses digital calibration

techniques that minimize offset and gain error.

Drift Considerations

The AD7714 uses chopper stabilization techniques to minimize

input offset drift. Charge injection in the analog switches and

dc leakage currents at the sampling node are the primary

sources of offset voltage drift in the converter. The dc input

leakage current is essentially independent of the selected gain.

Gain drift within the converter depends primarily upon the

temperature tracking of the internal capacitors. It is not af-

fected by leakage currents.

Measurement errors due to offset drift or gain drift can be elimi-

nated at any time by recalibrating the converter or by operating

the part in the background calibration mode. Using the system

calibration mode can also minimize offset and gain errors in the

signal conditioning circuitry. Integral and differential linearity

errors are not significantly affected by temperature changes.

POWER SUPPLIES

No specific power sequence is required for the AD7714; either

the AV

or the DV

supply can come up first. While the

latch-up performance of the AD7714 is good, it is important

that power is applied to the AD7714 before signals at REF␣ IN,

AIN or the logic input pins in order to avoid latch-up. If this is

not possible, then the current which flows in any of these pins

should be limited. If separate supplies are used for the AD7714

and the system digital circuitry, then the AD7714 should be

powered up first. If it is not possible to guarantee this, then

current limiting resistors should be placed in series with the

logic inputs to again limit the current.

Supply Current

The current consumption on the AD7714 is specified for sup-

plies in the range +3␣ V to +3.6␣ V and in the range +4.75␣ V to

+5.25␣ V. The part operates over a +2.85␣ V to +5.25␣ V supply

range and the I

for the part varies as the supply voltage varies

over this range. Figure 5 shows the variation of the typical

with V

voltage for both a 1 MHz external clock and a

2.4576 MHz external clock at +25°C. The AD7714 is operated

in unbuffered mode and the internal boost bit on the part is

turned off. The relationship shows that the I

is minimized by

operating the part with lower V

voltages. I

on the AD7714

is also minimized by using an external master clock or by opti-

mizing external components when using the on-chip oscillator

circuit. The Y grade part is specified from 2.7 V to 3.3 V and

4.75 V to 5.25 V.

AD7714

REV. C–28–

SUPPLY VOLTAGE (AVDD & DVDD) – Volts

2.85

0.9

1.0

0.8

0.7

0.6

0.5

0.4

0.3

0.2

0.1

3.15 3.45 4.05 4.35 4.65 4.95 5.253.75

MCLK IN = 2.4576MHz

MCLK IN = 1MHz

SUPPLY CURRENT (AVDD & DVDD) – mA

Figure 5. I

vs. Supply Voltage

Grounding and Layout

Since the analog inputs and reference input are differential,

most of the voltages in the analog modulator are common-mode

voltages. The excellent Common-Mode Rejection of the part

will remove common-mode noise on these inputs. The analog

and digital supplies to the AD7714 are independent and sepa-

rately pinned out to minimize coupling between the analog and

digital sections of the device. The digital filter will provide

rejection of broadband noise on the power supplies, except at

integer multiples of the modulator sampling frequency. The

digital filter also removes noise from the analog and reference

inputs provided those noise sources do not saturate the analog

modulator. As a result, the AD7714 is more immune to noise

interference that a conventional high resolution converter. How-

ever, because the resolution of the AD7714 is so high and the

noise levels from the AD7714 so low, care must be taken with

regard to grounding and layout.

The printed circuit board which houses the AD7714 should be

designed such that the analog and digital sections are separated

and confined to certain areas of the board. This facilitates the

use of ground planes which can be separated easily. A minimum

etch technique is generally best for ground planes as it gives the

best shielding. Digital and analog ground planes should only be

joined in one place. If the AD7714 is the only device requiring

an AGND to DGND connection, then the ground planes

should be connected at the AGND and DGND pins of the

AD7714. If the AD7714 is in a system where multiple devices

require AGND to DGND connections, the connection should

still be made at one point only, a star ground point which

should be established as close as possible to the AD7714.

Avoid running digital lines under the device as these will couple

noise onto the die. The analog ground plane should be allowed

to run under the AD7714 to avoid noise coupling. The power

supply lines to the AD7714 should use as large a trace as pos-

sible to provide low impedance paths and reduce the effects of

glitches on the power supply line. Fast switching signals like

clocks should be shielded with digital ground to avoid radiating

noise to other sections of the board and clock signals should

never be run near the analog inputs. Avoid crossover of digital

and analog signals. Traces on opposite sides of the board should

run at right angles to each other. This will reduce the effects of

feedthrough through the board. A microstrip technique is by far

the best but is not always possible with a double-sided board. In

this technique, the component side of the board is dedicated to

ground planes while signals are placed on the solder side.

Good decoupling is important when using high resolution

ADCs. All analog supplies should be decoupled with 10␣ µF

tantalum in parallel with 0.1␣ µF capacitors to AGND. To

achieve the best from these decoupling components, they have

to be placed as close as possible to the device, ideally right up

against the device. All logic chips should be decoupled with

0.1␣ µF disc ceramic capacitors to DGND. In systems where a

common supply voltage is used to drive both the AV

and

of the AD7714, it is recommended that the system’s

supply is used. This supply should have the recommended

analog supply decoupling capacitors between the AV

pin of

the AD7714 and AGND and the recommended digital supply

decoupling capacitor between the DV

pin of the AD7714 and

DGND.

Evaluating the AD7714 Performance

The recommended layout for the AD7714 is outlined in the

evaluation board for the AD7714. The evaluation board pack-

age includes a fully assembled and tested evaluation board,

documentation, software for controlling the board over the

printer port of a PC and software for analyzing the AD7714’s

performance on the PC. For the AD7714-5, the evaluation

board order number is EVAL-AD7714-5EB and for the AD7714-3,

the order number is EVAL-AD7714-3EB.

Noise levels in the signals applied to the AD7714 may also

affect performance of the part. The AD7714 allows two tech-

niques for evaluating the true performance of the part, indepen-

dent of the analog input signal. These schemes should be used

after a calibration has been performed on the part.

The first of these is to select the AIN6/AIN6 input channel

arrangement. In this case, the differential inputs to the AD7714

are internally shorted together to provide a zero differential

voltage for the analog modulator. External to the device, the

AIN6 input should be connected to a voltage that is within the

allowable common-mode range of the part.

The second scheme is to evaluate the part with a voltage near

the input full scale voltage for a gain of 1. To do this, the refer-

ence voltage for the part should be applied to the analog input.

This will give a fixed full-scale reading from the device. If the

zero-scale calibration coefficient is now read from the device,

increased by a number equivalent to about 200 decimal and this

value reloaded to the zero-scale calibration register, the input

range will be offset such that a voltage equal to reference voltage

no longer corresponds to a full-scale reading. This allows the

user to evaluate the noise performance of the part with a near

full-scale voltage.

AD7714

REV. C –29–

DIGITAL INTERFACE

The AD7714’s programmable functions are controlled using a

set of on-chip registers as previously outlined. Data is written to

these registers via the part’s serial interface, and read access to

the on-chip registers is also provided by this interface. All com-

munications to the part must start with a write operation to the

Communications Register. After power-on or RESET, the de-

vice expects a write to its Communications Register. The data

written to this register determines whether the next operation to

the part is a read or a write operation and also determines to

which register this read or write operation occurs. Therefore,

write access to any of the other registers on the part starts with a

write operation to the Communications Register followed by a

write to the selected register. A read operation from any register

on the part (including the output data register) starts with a

write operation to the Communications Register followed by a

read operation from the selected register.

The AD7714’s serial interface consists of five signals, CS,

SCLK, DIN, DOUT and DRDY. The DIN line is used for

transferring data into the on-chip registers while the DOUT line

is used for accessing data from the on-chip registers. SCLK is

the serial clock input for the device and all data transfers (either

on DIN or DOUT) take place with respect to this SCLK signal.

The DRDY line is used as a status signal to indicate when data

is ready to be read from the AD7714’s data register. DRDY

goes low when a new data word is available in the output regis-

ter. It is reset high when a read operation from the data register

is complete. It also goes high prior to the updating of the output

that a data read is not attempted while the register is being

updated. CS is used to select the device. It can be used to de-

code the AD7714 in systems where a number of parts are con-

nected to the serial bus.

The AD7714 serial interface can operate in three-wire mode by

tying the CS input low. In this case, the SCLK, DIN and

DOUT lines are used to communicate with the AD7714 and

the status of DRDY can be obtained by interrogating the MSB

of the Communications Register.

Figures 6 and 7 show timing diagrams for interfacing to the

AD7714 with CS used to decode the part. Figure 6 is for a read

operation from the AD7714’s output shift register, while Figure

7 shows a write operation to the input shift register. Both dia-

grams are for the POL input at a logic high; for operation with

the POL input at a logic low simply invert the SCLK waveform

shown in the diagrams. It is possible to read the same data

twice from the output register even though the DRDY line

returns high after the first read operation. Care must be taken,

however, to ensure that the read operations have been com-

pleted before the next output update is about to take place.

The serial interface can be reset by exercising the RESET input

on the part. It can also be reset by writing a series of 1s on the

DIN input. If a logic 1 is written to the AD7714 DIN line for at

least 32 serial clock cycles the serial interface is reset. This

ensures in three-wire systems that if the interface gets lost, either

via a software error or by some glitch in the system, it can be

reset back into a known state. This state returns the interface to

where the AD7714 is expecting a write operation to the Com-

munications Register. This operation does not in itself reset the

contents of any registers but since the interface was lost, the

information that was written to any of the registers is unknown

and it is advisable to set up all registers again.

Figure 6. Read Cycle Timing Diagram (POL = 1)

Figure 7. Write Cycle Timing Diagram (POL = 1)

DOUT

SCLK

DRDY

MSB

t5t7t9

LSB

t8t6

t3t10

DIN

SCLK

MSB

t12 t15

LSB

t16

t14

t11

t13

AD7714

REV. C–30–

Figure 8. Flowchart for Setting Up and Reading from the AD7714

CONFIGURING THE AD7714

The AD7714 contains eight on-chip registers that can be

accessed via the serial interface. Communication with any of

these registers is initiated by writing to the Communications

which is used to configure all registers after a power-up or reset.

The flowchart also shows two different read options—the first

where the DRDY pin is polled to determine when an update of

the data register has taken place, the second where the DRDY

bit of the Communications Register is interrogated to see if a

data register update has taken place. Also included in the flow-

ing diagram is a series of words which should be written to the

registers for a particular set of operating conditions. These con-

ditions are test channel (AIN6/AIN6), gain of 1, burnout cur-

rent off, no filter sync, bipolar mode, 24-bit word length, boost

off and maximum filter word (4000 decimal).

START

CONFIGURE & INITIALIZE µC/µP SERIAL PORT

POWER-ON/RESET FOR AD7714

WRITE TO COMMUNICATIONS REGISTER SETTING UP

CHANNEL & SETTING UP NEXT OPERATION TO BE A

WRITE TO THE FILTER HIGH REGISTER (27 HEX)

WRITE TO FILTER HIGH REGISTER SETTING

UP REQUIRED VALUES (4F HEX)

WRITE TO FILTER LOW REGISTER SETTING

UP REQUIRED VALUES (A0 HEX)

WRITE TO MODE REGISTER SETTING UP REQUIRED

VALUES & INITIATING A CALIBRATION (20 HEX)

WRITE TO COMMUNICATIONS REGISTER SETTING UP

SAME CHANNEL & SETTING UP NEXT OPERATION TO

BE A WRITE TO THE FILTER LOW REGISTER (37 HEX)

WRITE TO COMMUNICATIONS REGISTER SETTING UP

SAME CHANNEL & SETTING UP NEXT OPERATION TO

BE A WRITE TO THE MODE REGISTER (17 HEX)

POLL DRDY PIN

DRDY

LOW?

YES

WRITE TO COMMUNICATIONS REGISTER SETTING UP

SAME CHANNEL & SETTING UP NEXT OPERATION TO

BE A READ FROM THE DATA REGISTER (5F HEX)

READ FROM DATA REGISTER

DRDY

LOW?

POLL DRDY BIT OF COMMUNICATIONS REGISTER

YES

WRITE TO COMMUNICATIONS REGISTER SETTING UP

SAME CHANNEL & SETTING UP NEXT OPERATION TO

BE A READ FROM THE DATA REGISTER (5F HEX)

READ FROM DATA REGISTER

WRITE TO COMMUNICATIONS REGISTER SETTING UP

SAME CHANNEL & SETTING UP NEXT OPERATION TO BE A

READ FROM THE COMMUNICATIONS REGISTER (0F HEX)

READ FROM COMMUNICATIONS REGISTER

AD7714

REV. C –31–

The 68HC11 is configured in the master mode with its CPOL

bit set to a logic zero and its CPHA bit set to a logic one. When

the 68HC11 is configured like this, its SCLK line idles low

between data transfers. Therefore, the POL input of the AD7714

should be hard-wired low. For systems where it is preferable

that the SCLK idle high, the CPOL bit of the 68HC11 should

be set to a logic 1 and the POL input of the AD7714 should be

hard-wired to a logic high.

AD7714

DATA OUT

SCLK

SYNC

68HC11

DVDD

RESET

DATA IN

POL

SCK

MISO

MOSI

DVDD

Figure 9. AD7714 to 68HC11 Interface

The AD7714 is not capable of full duplex operation. If the

AD7714 is configured for a write operation, no data appears on

the DATA OUT lines even when the SCLK input is active.

Similarly, if the AD7714 is configured for a read operation, data

presented to the part on the DATA IN line is ignored even

when SCLK is active.

Coding for an interface between the 68HC11 and the AD7714

is given in Table XV. In this example, the DRDY output line of

the AD7714 is connected to the PC0 port bit of the 68HC11

and is polled to determine its status.

AD7714 to 8051 Interface

An interface circuit between the AD7714 and the 8XC51 mi-

crocontroller is shown in Figure 10. The diagram shows the

minimum number of interface connections with CS on the

AD7714 hard-wired low. In the case of the 8XC51 interface the

minimum number of interconnects is just two. In this scheme,

the DRDY bit of the Communications Register is monitored to

determine when the Data Register is updated. The alternative

scheme, which increases the number of interface lines to three,

is to monitor the DRDY output line from the AD7714. The

monitoring of the DRDY line can be done in two ways. First,

DRDY can be connected to one of the 8XC51’s port bits (such

as P1.0) which is configured as an input. This port bit is then

polled to determine the status of DRDY. The second scheme is

to use an interrupt driven system in which case, the DRDY

output is connected to the INT1 input of the 8XC51. For

MICROCOMPUTER/MICROPROCESSOR INTERFACING

The AD7714’s flexible serial interface allows for easy interface

to most microcomputers and microprocessors. The flowchart of

Figure 8 outlines the sequence which should be followed when

interfacing a microcontroller or microprocessor to the AD7714.

Figures 9, 10 and 11 show some typical interface circuits.

The serial interface on the AD7714 has the capability of operat-

ing from just three wires and is compatible with SPI interface

protocols. The three-wire operation makes the part ideal for

isolated systems where minimizing the number of interface lines

minimizes the number of opto-isolators required in the system.

The rise and fall times of the digital inputs to the AD7714

(especially the SCLK input) should be no longer than 1␣ µs.

Most of the registers on the AD7714 are 8-bit registers which

facilitates easy interfacing to the 8-bit serial ports of microcon-

trollers. Some of the registers on the part are up to 24 bits, but

data transfers to these 24-bit registers can consist of a full 24-bit

transfer or three 8-bit transfers to the serial port of the micro-

controller. DSP processors and microprocessors generally trans-

fer 16 bits of data in a serial data operation. Some of these

processors, such as the ADSP-2105, have the facility to program

the amount of cycles in a serial transfer. This allows the user to

tailor the number of bits in any transfer to match the register

length of the required register in the AD7714.

Even though some of the registers on the AD7714 are only eight

bits in length, communicating with two of these registers in

successive write operations can be handled as a single 16-bit

data transfer if required. For example, if the Mode Register is to

be updated, the processor must first write to the Communica-

tions Register (saying that the next operation is a write to the

Mode Register) and then write eight bits to the Mode Register.

This can all be done in a single 16-bit transfer if required be-

cause once the eight serial clocks of the write operation to the

Communications Register have been completed the part imme-

diately sets itself up for a write operation to the Mode Register.

AD7714 to 68HC11 Interface

Figure 9 shows an interface between the AD7714 and the

68HC11 microcontroller. The diagram shows the minimum

(three-wire) interface with CS on the AD7714 hard-wired low.

In this scheme, the DRDY bit of the Communications Register

is monitored to determine when the Data Register is updated.

An alternative scheme, which increases the number of interface

lines to four, is to monitor the DRDY output line from the

AD7714. The monitoring of the DRDY line can be done in two

ways. First, DRDY can be connected to one of the 68HC11’s

port bits (such as PC0) which is configured as an input. This

port bit is then polled to determine the status of DRDY. The

second scheme is to use an interrupt driven system in which

case, the DRDY output is connected to the IRQ input of the

68HC11. For interfaces which require control of the CS input

on the AD7714, one of the port bits of the 68HC11 (such as

PC1), which is configured as an output, can be used to drive the

CS input.

AD7714

REV. C–32–

interfaces which require control of the CS input on the AD7714,

one of the port bits of the 8XC51 (such as P1.1), which is con-

figured as an output, can be used to drive the CS input.

AD7714

DATA OUT

POL

SYNC

8XC51

RESET

DATA IN

SCLK

P3.0

P3.1

DVDD

Figure 10. AD7714 to 8051 Interface

The 8XC51 is configured in its Mode 0 serial interface mode.

Its serial interface contains a single data line. As a result, the

DATA OUT and DATA IN pins of the AD7714 should be

connected together. The serial clock on the 8XC51 idles high

between data transfers and, therefore, the POL input of the

AD7714 should be hard-wired to a logic high. The 8XC51

outputs the LSB first in a write operation while the AD7714

expects the MSB first so the data to be transmitted has to be

rearranged before being written to the output serial register.

Similarly, the AD7714 outputs the MSB first during a read

operation while the 8XC51 expects the LSB first. Therefore, the

data that is read into the serial buffer needs to be rearranged

before the correct data word from the AD7714 is available in

the accumulator.

AD7714 to ADSP-2103/ADSP-2105 Interface

Figure 11 shows an interface between the AD7714 and the

ADSP-2103/ADSP-2105 DSP processor. In the interface shown,

the DRDY bit of the Communications Register is again moni-

tored to determine when the Data Register is updated. The

alternative scheme is to use an interrupt driven system in which

case, the DRDY output is connected to the IRQ2 input of the

ADSP-2103/ADSP-2105. The RFS and TFS pins of the

ADSP-2103/ADSP-2105 are configured as active low outputs

and the ADSP-2103/ADSP-2105 serial clock line, SCLK, is

also configured as an output. The POL pin of the AD7714 is

hard-wired low. Because the SCLK from the ADSP-2103/

ADSP-2105 is a continuous clock, the CS of the AD7714 must

be used to gate off the clock once the transfer is complete. The

CS for the AD7714 is active when either the RFS or TFS

outputs from the ADSP-2103/ADSP-2105 are active. The serial

clock rate on the ADSP-2103/ADSP-2105 should be limited to

3␣ MHz to ensure correct operation with the AD7714.

AD7714

DATA OUT

SYNC

ADSP-2103/2105

RESET

DATA IN

SCLK

RFS

DVDD

TFS

POL

SCLK

Figure 11. AD7714 to ADSP-2103/ADSP-2105 Interface

CODE FOR SETTING UP THE AD7714

Table XV gives a set of read and write routines in C code for

interfacing the 68HC11 microcontroller to the AD7714. The

sample program sets up the various registers on the AD7714

and reads 1000 samples from the part into the 68HC11. The

setup conditions on the part are exactly the same as those out-

lined for the flowchart of Figure 8. In the example code given

here the DRDY output is polled to determine if a new valid

word is available in the output register.

The sequence of the events in this program are as follows:

1. Write to the Communications Register, setting the channel.

2. Write to the Filter High Register, setting the 4 MSBs of the

filter word and setting the part for 24-bit read, bipolar mode

with boost off.

3. Write to the Filter Low Register, setting the 8 LSBs of the

filter word.

4. Write to the Mode Register, setting the part for a gain of 1,

burnout current off, no filter synchronization and initiating a

self-calibration.

5. Poll the DRDY Output.

6. Read the data from the Data Register.

7. Loop around doing steps 5 and 6 until the specified number

of samples have been taken.

AD7714

REV. C –33–

Table XV. C Code for Interfacing AD7714 to 68HC11

/* This program has read and write routines for the 68HC11 to interface to the AD7714 and the sample

program sets the various registers and then reads 1000 samples from the part. */

#include <math.h>

#include <io6811.h>

#define NUM_SAMPLES 1000 /* change the number of data samples */

#define MAX_REG_LENGTH 3 /* this says that the max length of a register is 3 bytes */

Writetoreg (int);

Read (int,char);

char *datapointer = store;

char store[NUM_SAMPLES*MAX_REG_LENGTH + 30];

void main()

{/* the only pin that is programmed here from the 68HC11 is the /CS and this is why the PC2 bit

of PORTC is made as an output */

char a;

DDRC = 0x04; /* PC2 is an output the rest of the port bits are inputs */

PORTC | = 0x04; /* make the /CS line high */

Writetoreg(0x27); /* set the channel AIN6/AIN6 and set the next operation as write to the filter high

Writetoreg(0x4f); /* set Bipolar mode, 24 bits, boost off, all 4 MSBs of filterword to 1 */

Writetoreg(0x37); /* set the next operation as a write to the filter low register */

Writetoreg(0xA0); /* max filter word allowed for low part of the filterword */

Writetoreg(0x17); /* set the operation as a write to the mode register */

Writetoreg(0x20); /* set gain to 1, burnout current off, no filter sync, and do a self calibration */

while(PORTC & 0x10); /* wait for /DRDY to go low */

for(a=0;a<NUM_SAMPLES;a++);

{

Writetoreg(0x5f); /*set the next operation for 24 bit read from the data register */

Read(NUM_SAMPES,3);

}

Writetoreg(int byteword);

{

int q;

SPCR = 0x3f;

SPCR = 0X7f; /* this sets the WiredOR mode(DWOM=1), Master mode(MSTR=1), SCK idles high(CPOL=1), /SS

can be low always (CPHA=1), lowest clock speed(slowest speed which is master clock /32 */

DDRD = 0x18; /* SCK, MOSI outputs */

q = SPSR;

q = SPDR; /* the read of the staus register and of the data register is needed to clear the interrupt

which tells the user that the data transfer is complete */

PORTC &= 0xfb; /* /CS is low */

SPDR = byteword; /* put the byte into data register */

while(!(SPSR & 0x80)); /* wait for /DRDY to go low */

PORTC |= 0x4; /* /CS high */

}

Read(int amount, int reglength)

{

int q;

SPCR = 0x3f;

SPCR = 0x7f; /* clear the interupt */

DDRD = 0x10; /* MOSI output, MISO input, SCK output */

while(PORTC & 0x10); /* wait for /DRDY to go low */

PORTC & 0xfb ; /* /CS is low */

for(b=0;b<reglength;b++)

{

SPDR = 0;

while(!(SPSR & 0x80)); /* wait until port ready before reading */

*datapointer++=SPDR; /* read SPDR into store array via datapointer */

}

PORTC|=4; /* /CS is high */

}

AD7714

REV. C–34–

arranged in a bridge network and gives a differential output

voltage between its OUT(+) and OUT(–) terminals. With rated

full-scale pressure (in this case 300 mmHg) on the transducer,

the differential output voltage is 3 mV/Volt of the input voltage

(i.e., the voltage between its IN(+) and IN(–) terminals).

Assuming a 5 V excitation voltage, the full-scale output range

from the transducer is 15 mV. The excitation voltage for the

bridge is also used to generate the reference voltage for the

AD7714. Therefore, variations in the excitation voltage do not

introduce errors in the system. Choosing resistor values of

24␣ kΩ and 15 kΩ as per the diagram give a 1.92 V reference

voltage for the AD7714 when the excitation voltage is 5 V.

Using the part with a programmed gain of 128 results in the full

scale input span of the AD7714 being 15 mV which corresponds

with the output span from the transducer.

APPLICATIONS

The on-chip PGA allows the AD7714 to handle an analog input

voltage range as low as 10 mV full-scale with V

REF

= +1.25␣ V.

The differential inputs of the part allow this analog input range

to have an absolute value anywhere between AGND and AV

when the part is operated in unbuffered mode. It allows the user

to connect the transducer directly to the input of the AD7714.

The programmable gain front end on the AD7714 allows the

part to handle unipolar analog input ranges from 0 mV to

+20␣ mV to 0 V to +2.5␣ V and bipolar inputs of ±20 mV to

±2.5 V. Because the part operates from a single supply these

bipolar ranges are with respect to a biased-up differential input.

Pressure Measurement

One typical application of the AD7714 is pressure measure-

ment. Figure 12 shows the AD7714 used with a pressure trans-

ducer, the BP01 from Sensym. The pressure transducer is

CLOCK

GENERATION

OUT+

IN+

OUT–

IN– AUTO-ZEROED

MODULATOR

CHARGE BALANCING A/D

CONVERTER

DIGITAL

FILTER

AVDD

1mA

BUFFER

AVDD

SERIAL INTERFACE

STANDBY

SYNC

MCLK IN

MCLK OUT

1mA

REF IN (+)

REF IN (–)

AGND

DGND BUFFER DOUT DIN CS SCLK

POL

DRDY

RESET

24kV

15kV

DVDD

AD7714

AGND

+5V

EXCITATION VOLTAGE = +5V

AIN1

AIN2

AIN3

AIN4

AIN5

AIN6 A = 1–128

SWITCHING

MATRIX

PGA

Figure 12. Pressure Measurement Using the AD7714

AD7714

REV. C –35–

CLOCK

GENERATION

AUTO-ZEROED

MODULATOR

CHARGE BALANCING A/D

CONVERTER

DIGITAL

FILTER

AVDD

1mA

BUFFER

AVDD

SERIAL INTERFACE

STANDBY

SYNC

MCLK IN

MCLK OUT

1mA

REF IN (+)

REF IN (–)

AGND

DGND BUFFER DOUT DIN CS SCLK

POL

DRDY

RESET

DVDD

AD7714

DVDD

GND

+5V

AIN1

AIN2

AIN3

AIN4

AIN5

AIN6

VOUT

AD780

THERMOCOUPLE

JUNCTION

+5V

+VIN

A = 1–128

AGND

SWITCHING

MATRIX

PGA

Figure 13. Thermocouple Measurement Using the AD7714

Temperature Measurement

Another application area for the AD7714 is in temperature

measurement. Figure 13 outlines a connection from a thermo-

couple to the AD7714. In this application, the AD7714 is oper-

ated in its buffered mode to allow large decoupling capacitors

on the front end to eliminate any noise pickup which there may

have been in the thermocouple leads. When the AD7714 is

operated in buffered mode, it has a reduced common-mode

range. In order to place the differential voltage from the thermo-

couple on a suitable common-mode voltage, the AIN2 input of

the AD7714 is biased up at the reference voltage, +2.5␣ V.

AD7714

REV. C–36–

RTD Measurement

Figure 14 shows another temperature measurement application

for the AD7714. In this case, the transducer is an RTD (Resis-

tive Temperature Device), a PT100. The arrangement is a 4-

lead RTD configuration. There are voltage drops across the lead

resistances R

and R

but these simply shift the common-

mode voltage. There is no voltage drop across lead resistances

and R

as the input current to the AD7714 is very low

The

lead resistances present a small source impedance so it would

not generally be necessary to turn on the buffer on the AD7714.

If the buffer is required, the common-mode voltage should be

set accordingly by inserting a small resistance between the bot-

tom end of the RTD and AGND of the AD7714. In the appli-

cation shown an external 400␣ µA current source provides the

excitation current for the PT100 and it also generates the refer-

ence voltage for the AD7714 via the 6.25 kΩ resistor. Variations

in the excitation current do not affect the circuit as both the

input voltage and the reference voltage vary ratiometrically with

the excitation current. However, the 6.25␣ kΩ resistor must have

a low temperature coefficient to avoid errors in the reference

voltage over temperature.

CLOCK

GENERATION

AUTO-ZEROED

MODULATOR

CHARGE BALANCING A/D

CONVERTER

DIGITAL

FILTER

AVDD

1mA

BUFFER

AVDD

SERIAL INTERFACE

STANDBY

SYNC

MCLK IN

MCLK OUT

1mA

REF IN (+)

REF IN (–)

AGND

DGND

BUFFER DOUT DIN CS SCLK

POL

DRDY

RESET

DVDD

AD7714

+5V

400mA

AGND

RL4

RL3

RL2

RL1

RTD

6.25kV

AIN1

AIN2 A = 1–128

SWITCHING

MATRIX

PGA

Figure 14. RTD Measurement Using the AD7714

AD7714

REV. C –37–

Data Acquisition

The AD7714 with its three differential channels (or five pseudo-

differential channels) is suited to low bandwidth, high resolution

data acquisition systems. In addition, the three-wire digital

interface allows this data acquisition front end to be isolated

with just three optoisolators. The entire system can be operated

from a single +3␣ V or +5 V supply provided that the input sig-

nals to the AD7714’s analog inputs are all of positive polarity.

The low power operation of the AD7714 ensures that very little

power has to be brought across the isolation barrier. Figure 15

shows the AD7714 in an isolated data acquisition system.

CLOCK

GENERATION

CHARGE BALANCING A/D

CONVERTER

DIGITAL

FILTER

AVDD

1mA

BUFFER

AVDD

SERIAL INTERFACE

STANDBY

SYNC

MCLK IN

MCLK OUT

1mA

REF IN (+)

REF IN (–)

AGND

DGND BUFFER DOUT DIN CS SCLK

POL

DRDY

RESET

DVDD

AD7714

DVDD

GND

+5V

AIN1

AIN2

AIN3

AIN4

AIN5

AIN6

VOUT

AD780

+5V

+VIN

A = 1–128

AGND

IN1+

IN1–

IN2+

IN2–

IN3+

IN3–

MICROCONTROLLER OPTO-ISOLATORS

SWITCHING

MATRIX

PGA AUTO-ZEROED

MODULATOR

Figure 15. Data Acquisition System Using the AD7714

AD7714

REV. C–38–

Smart Transmitters

Another area where the low power, single supply, three-wire

interface capabilities is of benefit is in smart transmitters. Here,

the entire smart transmitter must operate from the 4␣ mA to

20␣ mA loop. Tolerances in the loop mean that the amount of

current available to power the transmitter is as low as 3.5␣ mA.

The AD7714 consumes only 500␣ µA, leaving 3␣ mA available for

the rest of the transmitter. Figure 16 shows a block diagram of a

smart transmitter which includes the AD7714. Not shown in

Figure 16 is the isolated power source required to power the

front end.

INPUT/OUTPUT

STAGE

SIGNAL

CONDITIONER

VOLTAGE

REFERENCE

MCLK

OUT

AD7714

ISOLATED SUPPLY

ISOLATION

BARRIER

SENSORS

RTD

ohm

TC LOOP

RTN

MCLK

IN COM

4–20mA

DVDD

AMBIENT

TEMP.

SENSOR

AVDD REF IN

AGND

DGND

ISOLATED GROUND

COM

VCC

3V VOLTAGE

REGULATOR

MAIN TRANSMITTER ASSEMBLY

BANDPASS

FILTER

D/A

CONVERTER

WAVEFORM

SHAPER

HART

MODEM

BELL 202

MICROCONTROLLER UNIT

*PID

*RANGE SETTING

*CALIBRATION

*LINEARIZATION

*OUTPUT CONTROL

*SERIAL COMMUNICATION

*HART PROTOCOL

VOLTAGE

REFERENCE

Figure 16. Smart Transmitter Using the AD7714

AD7714

REV. C –39–

PAGE INDEX

Topic Page

FEATURES . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

APPLICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

GENERAL DESCRIPTION . . . . . . . . . . . . . . . . . . . . . . . . . 1

PRODUCT HIGHLIGHTS . . . . . . . . . . . . . . . . . . . . . . . . . 1

AD7714-5 SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . 2

AD7714-3 SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . 3

SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

TIMING CHARACTERISTICS . . . . . . . . . . . . . . . . . . . . . 7

ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7

ABSOLUTE MAXIMUM RATINGS . . . . . . . . . . . . . . . . . 8

PIN CONFIGURATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . 8

PIN FUNCTION DESCRIPTION . . . . . . . . . . . . . . . . . . . 9

TERMINOLOGY . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10

AD7714-5 OUTPUT NOISE . . . . . . . . . . . . . . . . . . . . . . . 11

AD7714-3 OUTPUT NOISE . . . . . . . . . . . . . . . . . . . . . . . 12

BUFFERED MODE NOISE . . . . . . . . . . . . . . . . . . . . . . . 13

ON-CHIP REGISTERS . . . . . . . . . . . . . . . . . . . . . . . . . . . 14

Communications Register . . . . . . . . . . . . . . . . . . . . . . . . 14

Mode Register . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15

Filter Registers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17

Test Register . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

Data Register . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

Calibration Registers . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

CALIBRATION OPERATIONS . . . . . . . . . . . . . . . . . . . . 18

CIRCUIT DESCRIPTION . . . . . . . . . . . . . . . . . . . . . . . . 19

ANALOG INPUT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

Analog Input Ranges . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

Input Sample Rate . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

Burnout Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

Bipolar/Unipolar Inputs . . . . . . . . . . . . . . . . . . . . . . . . . . 21

REFERENCE INPUT . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

DIGITAL FILTERING . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

Filter Characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

Post-Filtering . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

ANALOG FILTERING . . . . . . . . . . . . . . . . . . . . . . . . . . . 23

CALIBRATION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23

Self-Calibration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23

System Calibration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24

System-Offset Calibration . . . . . . . . . . . . . . . . . . . . . . . . 25

Background Calibration . . . . . . . . . . . . . . . . . . . . . . . . . . 25

Span and Offset Limits . . . . . . . . . . . . . . . . . . . . . . . . . . 25

Power-Up and Calibration . . . . . . . . . . . . . . . . . . . . . . . . 26

USING THE AD7714 . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26

Clocking and Oscillator Circuit . . . . . . . . . . . . . . . . . . . . 26

System Synchronization . . . . . . . . . . . . . . . . . . . . . . . . . . 26

Reset Input . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27

Standby Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27

Accuracy . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25

Drift Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27

POWER SUPPLIES . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27

Supply Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27

Grounding and Layout . . . . . . . . . . . . . . . . . . . . . . . . . . . 28

Evaluating the AD7714 Performance . . . . . . . . . . . . . . . . 28

DIGITAL INTERFACE . . . . . . . . . . . . . . . . . . . . . . . . . . . 29

CONFIGURING THE AD7714 . . . . . . . . . . . . . . . . . . . . . 30

MICROCOMPUTER/MICROPROCESSOR

INTERFACING . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31

AD7714 to 68HC11 Interface . . . . . . . . . . . . . . . . . . . . . 31

AD7714 to 8051 Interface . . . . . . . . . . . . . . . . . . . . . . . . 31

AD7714 to ADSP-2103/ADSP-2105 Interface . . . . . . . . 32

Topic Page

CODE FOR SETTING UP AD7714 . . . . . . . . . . . . . . . . . 32

APPLICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34

Pressure Measurement . . . . . . . . . . . . . . . . . . . . . . . . . . . 34

Temperature Measurement . . . . . . . . . . . . . . . . . . . . . . . 35

RTD Measurement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36

Data Acquisition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37

Smart Transmitters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38

OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . 40

TABLE INDEX

Table Title Page

Table Ia. AD7714-5 Output Noise/Resolution vs. Gain

and First Notch for f

CLK IN

= 2.4576 MHz . . . 11

Table Ib. AD7714-5 Output Noise/Resolution vs. Gain

and First Notch for f

CLK IN

= 1 MHz . . . . . . . . 11

Table IIa. AD7714-3 Output Noise/Resolution vs. Gain

and First Notch for f

CLK IN

= 2.4576 MHz . . . . 12

Table IIb. AD7714-3 Output Noise/Resolution vs. Gain

and First Notch for f

CLK IN

= 1 MHz . . . . . . . . 12

Table III. AD7714-5 Buffered Mode Output Noise/

Resolution . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

Table IV. AD7714-3 Buffered Mode Output Noise/

Resolution . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

Table V. Communications Register . . . . . . . . . . . . . . . . 14

Table VI. Register Selection . . . . . . . . . . . . . . . . . . . . . . 14

Table VII. Channel Selection . . . . . . . . . . . . . . . . . . . . . . 15

Table VIII. Mode Register . . . . . . . . . . . . . . . . . . . . . . . . . 15

Table IX. Filter High Register . . . . . . . . . . . . . . . . . . . . . 17

Table X. Filter Low Register . . . . . . . . . . . . . . . . . . . . . 17

Table XI. Calibration Operations . . . . . . . . . . . . . . . . . . 18

Table XII. External R, C for No 16-Bit Gain Error . . . . . 20

Table XIII. External R, C for No 20-Bit Gain Error . . . . . 20

Table XIV. Input Sampling Frequency vs. Gain . . . . . . . . 21

Table XV. C Code for AD7714 to 68HC11 Interface . . . 33

AD7714

REV. C–40–

PRINTED IN U.S.A. C1972a–0–6/98

OUTLINE DIMENSIONS

Dimensions are shown in inches and (mm).

24-Lead Plastic DIP

(N-24)

112

13 0.260 ± 0.001

(6.61 ± 0.03)

PIN 1

1.228 (31.19)

1.226 (31.14)

0.11 (2.79)

0.016 (2.28) SEATING

PLANE

0.02 (0.5)

0.019 (0.41)

0.130 (3.30)

0.128 (7.62)

0.07 (1.78)

0.05 (1.27)

0.32 (8.128)

0.30 (7.62)

0.011 (0.28)

0.009 (0.23)

158

NOTES:

1. LEAD NO. 1 IDENTIFIED BY DOT OR NOTCH

2. PLASTIC LEADS WILL BE EITHER SOLDER DIPPED OR TIN PLATED

IN ACCORDANCE WITH MIL-M-38510 REQUIREMENTS.

24-Lead Wide Body SOIC

(R-24)

24 13

0.608 (15.45)

0.596 (15.13)

0.414 (10.52)

0.398 (10.10)

0.299 (7.6)

0.291 (7.39)

PIN 1

SEATING

PLANE

0.01 (0.254)

0.006 (0.15) 0.019 (0.49)

0.014 (0.35)

0.096 (204)

0.089 (2.26)

0.05 (1.27)

BSC 0.013 (0.32)

0.009 (0.23)

0.0500 (1.27)

0.0157 (0.40)

0.03 (0.76)

0.02 (0.51)

NOTES:

1. LEAD NO. 1 IDENTIFIED BY DOT.

2. SOIC LEADS WILL BE EITHER TIN PLATED OR SOLDER DIPPED

IN ACCORDANCE WITH MIL-M-38510 REQUIREMENTS.

28-Lead Shrink Small Outline Package SSOP

(RS-28)

0.009 (0.229)

0.005 (0.127)

0.03 (0.762)

0.022 (0.558)

0.008 (0.203)

0.002 (0.050)

0.07 (1.79)

0.066 (1.67)

SEATING

PLANE

0.0256 (0.65)

BSC

0.311 (7.9)

0.301 (7.64)

0.212 (5.38)

0.205 (5.207)

28 15

141

0.407 (10.34)

0.397 (10.08)

PIN 1

NOTES:

1. LEAD NO. 1 IDENTIFIED BY DOT.

2. LEADS WILL BE EITHER TIN PLATED OR SOLDER DIPPED

IN ACCORDANCE WITH MIL-M-38510 REQUIREMENTS.

24-Lead Thin Shrink Small Outline Package TSSOP

(RU-24)

24 13

0.311 (7.90)

0.303 (7.70)

0.256 (6.50)

0.246 (6.25)

0.177 (4.50)

0.169 (4.30)

PIN 1

SEATING

PLANE

0.006 (0.15)

0.002 (0.05)

0.0118 (0.30)

0.0075 (0.19)

0.0256 (0.65)

BSC

0.0433

(1.10)

MAX

0.0079 (0.20)

0.0035 (0.090)

0.028 (0.70)

0.020 (0.50)